Analysis of Scalable Resonant DC–DC Converter Using GaN Switches for xEV Charging Stations

Abstract

In this research, an innovative electric vehicle (EV) charger is designed and presented for xEV charging stations. The key feature of our system is a scalable, interleaved inductor–inductor–capacitor (iL²C) DC-DC converter operation. The proposed system employs two parallel L²C converters with 8-GaN switches on the primary side and a shared rectifier circuit on the secondary side. This configuration not only amplifies the resonant tank internal currents and losses generated by the switches but also improves current sharing. A novel closed-loop technique is proposed with a constant-voltage method of operation, along with a hybrid control scheme of variable frequency + phase shift modulation (VFPSM). To examine the controller and converter’s performance, an experimental demonstration is conducted under varying load conditions, including full load, half load, and light load, where the source voltage and load voltage are maintained at constant levels of 400 V_in and 48 V₀, respectively. Furthermore, line regulation is conducted and verified to accommodate a broad input voltage range of 300 V_in–500 V_in and 500 V_in–300 V_in while maintaining an output voltage of 48 V₀ at 3.3 kW, 1.65 kW, and 0.33 kW with a peak efficiency of 98.2%.

1. Introduction

Due to the contamination caused by toxic gases, carbon emissions, and unpredictable weather patterns, conventional internal combustion engines are approaching their maximum capabilities. There is a global initiative to implement environmentally friendly transportation in order to mitigate the effects of climate change. Electric vehicles (EVs) are increasingly becoming popular due to their environmentally advantageous nature, aiming to replace internal combustion engines (ICEs) and reduce carbon dioxide emissions [1,2,3]. By transitioning to electric vehicles, such as battery electric vehicles (BEVs), fuel cell electric vehicles (FCEVs), hybrid electric vehicles (HEVs), ultra-capacitor electric vehicles (UCEVs), solar-charged electric vehicles (SCEVs), and plug-in hybrid electric vehicles (PHEVs), we can significantly reduce pollutants [4,5]. The goal of achieving a carbon-free society by 2030 in Europe and 2050 in India is ambitious but achievable with the swift adoption of EVs and the development of robust charging infrastructure. This transition not only contributes to environmental preservation but also promotes technological advancement and energy independence. The increasing demand for electric vehicles necessitates the development of a strong and efficient charging infrastructure. The information in [6,7] covers power electronics for several applications such as solar energy, fuel cells, renewable energy, and electric cars. It also discusses optimization approaches for various charging topologies. Figure 1 illustrates a standard block diagram of a charging infrastructure.

In the present scenario, the EV industry is looking for a scalable design that can operate over wide operating ranges and easily adjust to different power requirements [8]. The need for scalability is increased because battery charging technology has improved over a wide range from 48 V to 800 V and even larger in some cases. The scalable designs are helpful with their quick adaptability of operation to the desired voltage range [9]. Resonant converters are more popular for their merit of operation over wide operating ranges; hence, this paper deals with an advanced resonant converter named the iL²C converter, which is an extension of the L²C converter [10,11].

According to our understanding, L²C resonant converters are gaining popularity in comparison to numerous traditional DC-DC converters [12,13]. This is due to their inherent benefits, including primary-side zero voltage switching (ZVS) operation across all load circumstances and secondary-side zero current switching (ZCS) operation with synchronous rectification [14,15]. The ZVS function can be compromised if the switching frequency, denoted as f_s, is too low. Consequently, designers must take this into account when dealing with applications that require a wide voltage gain [16,17]. Furthermore, the use of the L²C converter as an EV charger is subject to many limitations. For instance, augmenting the load capacity at a high power density results in amplified voltage and current fluctuations, which can potentially harm the product [18,19]. The volume of the resonant tank is limited, resulting in increased circulating current and reduced efficiency [20].

This work introduces an iL²C converter as an electric vehicle charger to address the research gap in L²C converters [21]. In order to increase the load capacity, this converter employs two L²C converters operating in parallel. Consequently, the power losses in the transformer are reduced during high-current applications. Over time, numerous control techniques have been suggested to enhance soft switching, ensure precise regulation, and minimize the circulating current [22,23]. All of these control methods can be employed in different working areas of the converter, such as when the switching frequency (f_s) is less than the resonant frequency (f_r), when f_s is equal to f_r, and when f_s is more than f_r. These methods have an impact on the regulation of the converter. This article presents a hybrid control technique called the variable frequency + phase shift modulation (VFPSM) scheme. The strategy operates at a specified current value in either the zero current switching (ZCS) or zero voltage switching (ZVS) operating area. Furthermore, the iL²C DC-DC converter utilizes wideband gap semiconductor-based E-mode (enhancement mode) GaN technology.

The research contributions of this paper are as follows:

• The modeling of a scalable iL²C resonant converter with operation is presented.
• A hybrid control strategy is suggested and derived to improve the regulations.
• Simulations are presented for various load and line regulating conditions to check the converter performances.
• The experimental setup is demonstrated using E-mode GaN technology at 3.3 kW as an EV charger with an efficiency of 98.2%.

2. Modeling and Operation of iL²C Converter Topology

Figure 2 illustrates the resonant DC-DC topology of the iL²C, while Figure 3 depicts identical waveforms. It consists of two resonant circuits, each with a switching topology that incorporates a primary and secondary rectifier circuit utilizing a single diode bridge. It functions by employing a phase angle of 180° between the gate signals of two switching circuits. Equation (1) denotes that the resonant frequency of the converter is 150 kHz [21]. Additionally, the functioning of two switching topologies is comparable throughout their entire operational cycle; thus, the equations are examined in relation to the resonant network 1 and converter 1 [21,22]. The graphical representation of the gain performances of the converter is shown in Figure 4.

$$f_r=\frac{1}{2\pi \sqrt{L_r\ast C_r}}$$

(1)

Stage 1: All secondary-side diodes are deactivated prior to time zero, while switches S₂ and S₃ are in the active state. During the time interval t₀–t₁, the switches S₂ and S₃ are deactivated, while the body diodes S₁ and S₄ are activated using a negative resonant current of i_Lr1.

Stage 2: The voltage throughout transformer 1 T₁ is equivalent to the input voltage V_in between time points t₁ and t₂, and the switches S₁, S₄ and diodes D₁, D₄ are all in the ON state. Concurrently, the additional current of i_NS1 conducts via the filter capacitor C₀, while the resonant inductor L_r1 begins to circulate through the resonant capacitor C_r1. The voltage of the magnetizing inductance L_m1 is specified by Equation (2), and the current increases incrementally. Furthermore, the voltage across V_NS1 from time t₁ to time t₂ is equivalent to the output voltage V₀, as calculated in Equation (3), and subsequently, the voltage at the resonant tank is described in Equation (4) [24].

$$L_{m1}=\frac{V_0}{n}$$

(2)

$$V_{NS1}=V_0$$

(3)

$$\left\{\begin{array}{c}{i}_{Lr1}\left(t\right)=I_0\cos \left({\omega }_{r}t\right)+\frac{\left(n{V}_{in}-V_0-n{V}_0\right)}{n{Z}_r}\sin \left({\omega }_{r}t\right)\\ V_{Cr1}\left(t\right)=I_0{Z}_r\sin \left({\omega }_{r}t\right)+\left(V_0+\frac{V_0}{n}-V_{in}\right)\cos \left({\omega }_{r}t\right)\\ +V_{in}-\frac{V_0-V_{C0}}{n}\end{array}\right.$$

(4)

Stage 3: The switches S₂ and S₃ are activated at intervals t₂ through t₃, resulting in a positive current flow at i_Lr1, while the secondary-side diode rectifier circuit remains in an OFF state.

Stage 4: During the time intervals t₃ to t₄, the secondary-side diodes D₂ and D₄ are in the ON position in relation to the switches S₂ and S₃, because of the negative current circulation in i_NS1. Consequently, the secondary-side voltage of the transformer is −V_C0, which is exceedingly low, and the current i_Lm1 progressively diminishes. Conduction between the resonant inductor L_r1 and the resonant capacitor C_r1 commences simultaneously with the secondary current of i_NS1 charging the filter capacitor C₀. Equation (5) represents the voltage V_NS1 at this juncture, while Equation (6) depicts the state space equation [24,25,26].

$$V_{NS1}=-V_0$$

(5)

$$\left\{\begin{array}{c}{i}_{Lr1}\left(t\right)=I_{Lr1}\left(t_2\right)cos\left({\omega }_{r}t\right)+\frac{\left(n{V}_{in}-V_0-n{V}_{Cr1}(t_2\right)}{n{Z}_r}\mathit{sin}\left({\omega }_{r}t\right)\\ V_{Cr1}\left(t\right)=I_{Lr1}\left(t_2\right)Z_r\mathit{sin}\left({\omega }_{r}t\right)+\left(V_{Cr1}\left(t_2\right)-\frac{V_0}{n}-V_{in}\right)cos\left({\omega }_{r}t\right)\\ +V_{in}+\frac{V_0}{n}\end{array}\right.$$

(6)

Stage 5: From time t₄ to time t₅, the components D₂ and D₃ are both in the ON state, while the secondary-side transformer current i_NS1 is reduced to zero. Equation (7) represents the secondary side of the transformer voltage; consequently, the magnetizing inductance drops gradually [27,28]. Additionally, resonance occurs between the resonant capacitor C_r1 and the resonant inductor L_r1.

$$V_{NS1}=-V_0$$

(7)

The state space equations for stage 5 are defined in Equation (8) below:

$$\left\{\begin{array}{c}{i}_{Lr1}\left(t\right)=I_{Lr1}\left(t_4\right)cos\left({\omega }_{r}t\right)+\frac{\left(n{V}_{in}+V_0-n{V}_{C{r}_1}\left(t_4\right)\right.}{n{Z}_r}sin\left({\omega }_{r}t\right)\\ V_{Cr1}\left(t\right)=I_{Lr1}\left(t_4\right)Z_{r}sin\left({\omega }_{r}t\right)+\left(V_{Cr1}\left(t_4\right)-\frac{V_0}{n}-V_{in}\right)cos\left({\omega }_{r}t\right)\\ +V_{in}+\frac{V_0}{n}\end{array}\right.$$

(8)

Stage 6: From time t₅ to time t₆, the current i_NS1 on the secondary side of the transformer is entirely negative, accompanied by the absence of any diodes. The magnetizing inductance i_Lm1 begins to resonate with the resonant capacitor C_r1 and inductor L_r1 at this time.

The critical parameters of the iL²C converter for every single component are mentioned in [21,22,23,24]; the nominal input voltage is 400 V, and the output voltage is 48 V at a power range of 3.3 kW and a filter capacitor value of 100 µF, respectively. The associated step-by-step engineering calculations then followed with the appropriate values.

Equations (9)–(11) were utilized to derive a transformer T₁ turn ratio of 8.82, as well as the maximum and minimum values of voltage gain of the iL²C converter under the specified operating conditions.

$$n=\frac{V_{in\_min}}{V_{0\_min}}$$

(9)

$$M_{min}=\frac{n\ast V_{0\_min}}{V_{in\_max}}$$

(10)

$$M_{max}=\frac{n\ast V_{0\_max}}{V_{in\_min}}$$

(11)

where n is transformer turn ratio, V_{in_max} and V_{in_min} are maximum and minimum input voltages, V_{0_max} and V_{0_min} are maximum and minimum output voltages, followed by maximum and minimum voltage gain M_max and M_min [24].

The resonant tank design of the resonant inductance L_r1 was 22.11 µH; the capacitance C_r1 was 50.94 nF; the impedance Z₀₁ and the magnetizing inductance L_m1 were 20.83 Ω and 23 µH respectively with an equivalent resistance R_ac1 of 22.12 Ω and 55.81 Ω as the minimum and maximum values. The formulas are mentioned in Equations (12)–(16) [29,30].

$$R_{ac1}=\frac{8n^2}{{\pi }^2}\ast R_L$$

(12)

$$L_{m1}=\frac{n^2}{f_r}\ast \frac{V_{o\_crit}}{4\ast n\ast I_{sc}+\left[{\pi }^2\ast L\ast M_{crit}-4\right]\ast I_{o\_crit}}$$

(13)

$$L_{r1}=L\ast L_{m1}$$

(14)

$$C_{r1}=\frac{1}{L_{r1}\ast {\left(2\pi f_r\right)}^2}$$

(15)

$$Z_{01}=\sqrt{\frac{L_{r1}}{C_{r1}}}$$

(16)

3. Control Strategy

There are now multiple control approaches being studied for resonant converters mainly based on fixed-or variable-frequency techniques and phase shifting strategies. Nevertheless, in practical use, each of them possesses distinct constraints when it comes to converters operating within a wider range of input and output voltages. Typically, frequency modulation is effective but necessitates intricate magnetic construction, whereas the phase shifting approach is straightforward to build but challenging for a wide variety of input voltages [24]. This paper presents a hybrid control methodology, known as the VFPSM method, for an iL²C converter. The VFPSM method ensures consistent control strategy performance throughout all areas.

The switching frequency f_s is determined by comparing the output voltage V₀ to the reference input voltage V_ref, relying on the provided error voltage. The voltage control oscillator (VCO) can be utilized to compare the input and output voltage values [22,23,24], while also driving the gate signals. Figure 5 displays the control circuit. The controller regulates the output voltage V₀ and modulates the current, referred to as the CV mode of operation. The third-order transfer function, specified in Equation (17) with sub-equations in Equations (18)–(22), is valuable for constructing the outer feedback compensator [31,32,33].

$$Gvf\left(s\right)=\frac{V{\left(s\right)}_0}{w{\left(s\right)}_s}=G_{dc}\frac{X_{eq}^2+R_{eq}^2}{\left(s^2{L}_e^2+s{L}_e{R}_{eq}+X_{eq}^2\right)\left(1+s{R}_L{C}_f\right)+R_{eq}\left(s{L}_e+R_{eq}\right)}$$

(17)

$$L_e=L_r+\frac{1}{C_r{\mathsf{\Omega }}_s^2}=L_r\left(1+\frac{{\mathsf{\Omega }}_0^2}{{\mathsf{\Omega }}_s^2}\right)$$

(18)

$$R_{eq}=\frac{8}{{\pi }^2}{n}^2{R}_L$$

(19)

$$X_{eq}={\mathsf{\Omega }}_s{L}_r-\frac{1}{{\mathsf{\Omega }}_s{C}_r}$$

(20)

$$L_n=\frac{L_m}{L_r}$$

(21)

$$Q=\frac{\sqrt{\frac{L_r}{C_r}}}{n^2{R}_L}$$

(22)

The development of the switching frequency design was concluded based on the information provided in Figure 5. This design enables the generation of pulses that may be used to switch the switches ON and OFF [34,35]. The 180-degree phase shift technique is employed to control the input voltage within the wide range of 300 V_in–500 V_in, while the VFPSM methodology ensures precise regulation at both the line and load sides [36]. The operation of converters 1 and 2 involves a switching process that utilizes the phase shifting approach known as “before 180°” and “after 180°”, respectively, as seen in Figure 6 [37,38].

4. Simulation and Experimental Validation

MATLAB/Simulink was utilized to conduct the simulation analysis of the iL²C converter employing the VFPSM control technique. The essential design variables are enumerated in [24]. The evaluation of the converter is provided for various case studies. In the first instance, the nominal input and output voltage under full load, half load, and light load conditions were examined. The nominal input voltage of 400 V_in is depicted as a constant in Figure 7.

The outcomes of the simulated converter were assessed under controlled nominal load conditions of 68.75 A and 48 V₀ at maximum load (3.3 kW). Figure 8 shows the resonant tank waveforms of converter 1 and converter 2, and the converter load voltage and current waveforms are presented in Figure 9. The voltage and current ripple measurements indicated a deviation of +0.5 V (+1.04%) at a rated output of 48 V₀ and approximately +0.65 A (+0.94%) at a full load current of 68.75 A, respectively.

Figure 10 shows the resonant tank waveforms of converter 1 and converter 2; Figure 11 presents the simulated results for the half load condition, which is at 1.65 kW, 48 V, and 34.375 A, respectively. At an output of 48 V₀, both ripple voltages and currents were determined to have a voltage variance of +0.3 V (+0.62%) and a current variance of approximately +0.175 A (+0.5%) at a half load current of 38.375 A.

Figure 12 shows the resonant tank waveforms of converter 1 and converter 2; Figure 13 presents the simulated results at a light load which is 10% of the rated load, i.e., 0.33 kW, 48 V, and 6.875 A, respectively. At an output of 48 V₀, both ripple voltages and currents were determined to have a voltage variance of +0.15 V (+0.31%) and a current variance of approximately +0.017 A (+0.25%) at a half load current of 6.875 A.

4.1. Dynamic Performance Analysis of iL²C Converter for 300 V_in–500 V_in under Various Load Conditions

In this section, the dynamic performance of the iL²C converter with a variable input voltage of 300–500 V_in and an output voltage that stays constant at 48 V₀ is evaluated. The voltage increased abruptly from 300 to 500 V_in at 0.05 s, as depicted in Figure 14. Figure 15 illustrates the load voltage of 48 V and current waveforms at 3.3 kW. The sudden variation in load voltage and current took place at 0.05 s; the enlarged figures are also provided, and the settling time was within 1.1 ms.

Figure 16 illustrates the load voltage of 48 V and current waveforms at 1.65 kW. The sudden variation in load voltage and current took place at 0.05 s; the enlarged figures are also provided, and the settling time was within 1.0 ms.

Figure 17 illustrates the load voltage of 48 V and current waveforms at 0.33 kW. The sudden variation in load voltage and current took place at 0.05 s; the enlarged figures are also provided, and the settling time was within 1.3 ms.

4.2. Dynamic Performance Analysis of iL²C Converter for 500 V_in–300 V_in under Various Load Conditions

In this section, the dynamic performance of the iL²C converter with a variable input voltage of 500–300 V_in and an output voltage that stays constant at 48 V₀ is evaluated. The voltage increased abruptly from 500 to 300 V_in at 0.05 s, as depicted in Figure 18. Figure 19 illustrates the load voltage of 48 V and current waveforms at 3.3 kW. The sudden variation in load voltage and current took place at 0.05 s; the enlarged figures are also provided, and the settling time was within 1.2 ms.

Figure 20 illustrates the load voltage of 48 V and current waveforms at 1.65 kW. The sudden variation in load voltage and current took place at 0.05 s; the enlarged figures are also provided, and the settling time was within 1.2 ms.

Figure 21 illustrates the load voltage of 48 V and current waveforms at 0.33 kW. The sudden variation in load voltage and current took place at 0.05 s; the enlarged figures are also provided, and the settling time was within 1.4 ms.

4.3. Experimental Validation of iL²C Converter

The iL²C converter concept is constructed utilizing GaN technology and operates at a maximum rated load capacity of 3.3 kW. The purpose of this prototype is to demonstrate the performance line and load regulation of the converter. The range of input voltages is set between 300 V_in and 500 V_in, while the rated load voltage is maintained at 48 V₀. The converter’s effectiveness is evaluated under different load conditions [39,40]. The major side of the converter includes 8-GaN switches, specifically 4 in converter 1 and 4 in converter 2.

The production specifications of the GaN switch GS66508T are listed in [24], where the critical parameters are a drain source voltage of 650 V, a drain current of 30 A, and a low drain resistance of 50 mΩ, respectively. The E-mode GaN transistor possesses advanced characteristics such as an upper cooled technology, resulting in reduced junction–case thermal resistance at high power density [41,42]. It also exhibits decreased drain resistance, reverse recovery current, and zero reverse recovery losses. Additionally, it showcases well-designed gate charge characteristics [43]. The TMS320F28335 serves as the primary controller for the iL²C converter, functioning as a digital signal processor [44,45]. The prototype was constructed according to specific modeling parameters, including a magnetizing inductance of 23 µH for L_m1 and L_m2, resonant inductors of 21 µH for L_r1 and L_r2, resonant capacitors of 56 nF for C_r1 and C_r2, and a filter capacitor of 100 µF for C₀. The experimental setup is illustrated in Figure 22.

Figure 23 displays the empirical waveforms of the drain and gate voltages of a switch operating with zero voltage switching (ZVS). This clearly demonstrates the successful attainment of ZVS functioning using soft switching.

The load regulation of the converter was analyzed and is depicted in Figure 24 under different load situations of 3.3 kW, 1.65 kW, and 0.33 kW. The input voltage was kept constant at 400 V_in, while the output voltage was maintained at 48 V₀. Based on the findings, it was observed that the output voltage remained steady during load operations involving step-up and step-down conversion; this indicates effective voltage control. The findings were examined with a time scale of 100 ms for each division and a load voltage scale of 20 volts per division, shown by a blue line. The load current scale was set at 50 amperes per division, represented by a pink line. The input source voltage scale was set at 300 volts per division, represented by a red line.

As seen in Figure 25, the iL²C converter was validated for 3.3 kW at 48 V₀ and 68.75 A during variable input voltage conditions. The findings were examined with a time scale of 100 ms for each division and a load voltage scale of 20 volts per division, shown by an orange line. The load current scale was set at 50 amperes per division, represented by a pink line. The input source voltage scale was set at 300 volts per division, represented by a blue line. The dynamic responses at the line side were executed by introducing an abrupt alteration in input voltage at 0.05 s (500 ms), resulting in a little modification in load voltage. The corresponding current was then monitored and recorded, as depicted in Figure 25. The converter and controller demonstrated conformity with the theoretical and simulated studies.

The efficiency of the GaN-powered iL²C converter was evaluated at the recommended voltage settings of 400 V_in and 48 V₀. The efficiency was measured at various load conditions, specifically 3.3 kW, 3.0 kW, 2.5 kW, 2.0 kW, 1.65 kW, 1.0 kW, 0.5 kW, and 0.33 kW, sequentially. The efficiency was 98.2% at a power output of 3.3 kW, 98.7% at a power output of 1.65 kW, and 99.1% at a power output of 0.33 kW. Figure 26 displays the efficiency curve in relation to power.

5. Performance Analysis and Discussion

This section examines the steady-state and dynamic performance assessment of the converter in relation to line regulation and load regulation. The comparative assessment of simulation and experimental analysis of the converter was carried out during various conditions presented in Section 4. Moreover, the converter’s merits are discussed under various parameters at the end of this section.

5.1. Steady-State Performance Analysis

This analysis was conducted at constant input and load voltages, while varying the load. The voltage deviation and ripple percentage were measured to be +0.5 V (+1.04%), +0.3 V (+0.62%), and +0.15 V (+0.31%) at power levels of 3.3 kW, 1.65 kW, and 0.33 kW, respectively. Figure 27 and Figure 28 show the steady-state analysis under simulating conditions of load voltage deviation (LVD), average voltage ripple (%AVR), load current deviation (LCD), and average current ripple (%ACR), respectively [24]. The present discrepancy and fluctuation percentages were measured to be +0.65 A (+0.94%), +0.175 A (+0.5%), and +0.017 A (+0.25%) at power levels of 3.3 kW, 1.65 kW, and 0.33 kW, correspondingly.

The voltage variation and ripple percentage were measured to be +0.9 V (+1.87%), +0.7 V (+1.45%), and +0.52 V (+1.09%) for power levels of 3.3 kW, 1.65 kW, and 0.33 kW, respectively. Figure 29 and Figure 30 show the steady-state analysis under experimental conditions of LVD, %AVR, LCD, and %ACR, respectively [24]. The present difference and fluctuation percentages were measured as +1.26 A (+1.83%), +0.525 A (+1.52%), and +0.095 A (+1.23%) at power levels of 3.3 kW, 1.65 kW, and 0.33 kW, respectively.

5.2. Transient Performance Analysis

Table 1. Transient analysis under line regulation during simulations.

Operation Attributes		Variable Input Voltage	Max. LVD	% ALVR	Transient Analysis
					Voltage Dip	Avg % of Dip	Settling Time
Load Variation	3.3 kW	300 V–500 V	+0.40 V	+0.830%	+5.2 V	+10.83%	1.1 ms
		500 V–300 V	+0.42 V	+0.875%	−3.5 V	−7.29%	1.2 ms
	1.65 kW	300 V–500 V	+0.38 V	+0.791%	+4.8 V	+10.00%	1.0 ms
		500 V–300 V	+0.40 V	+0.833%	−5.8 V	−12.08%	1.2 ms
	0.33 kW	300 V–500 V	+0.37 V	+0.770%	+6.35 V	+13.23%	1.33 ms
		500 V–300 V	+0.38 V	+0.791%	−8.70 V	−18.12%	1.40 ms

shows the transient analysis during simulation conditions of the load voltage deviations, %avg load ripples, transient dip, and %avg dip of the voltage during line regulation (i.e., 300 V_in–500 V_in and 500 V_in–300 V_in) tabulated by maintaining the load voltage at 48 V₀ and the load current at 68.75 A, 38.375 A, and 6.875 A, respectively. When the load was 3.3 kW, for (300 V_in–500 V_in) and (500 V_in–300 V_in), the max LVD was found to be +0.40 V and +0.42 V, and the %ALVR was +0.830% and +0.875%, respectively. The transient voltage dip was found to be +5.2 V, −3.5 V, and its average percentage dip was +10.83%, −7.29%, respectively, with a settling time of 1.1 ms and 1.2 ms. In the case of 1.65 kW, the max LVD was found to be +0.38 V and +0.40 V, and the %ALVR was +0.791% and +0.833%, respectively. The transient voltage dip was found to be +4.8 V, −5.8 V, and its average percentage dip was +10.00%, −12.08%, respectively, with a settling time of 1.0 ms and 1.2 ms. Lastly, in the case of 0.33 kW, the max LVD was found to be +0.37 V and +0.38 V, and the %ALVR was +0.770% and +0.791%, respectively. The transient voltage dip was found to be +6.35 V, −8.70 V, and its average percentage dip was +13.23%, −18.12%, respectively, with a settling time of 1.33 ms and 1.4 ms.

Table 2. Transient analysis of load current under line regulation during simulation.

Operation Attributes		Variable Input Voltage	Max. LCD	% ALCR	Transient Analysis
					Current Dip	Avg % of Dip	Settling Time
Load Variation	3.3 kW	300 V–500 V	+0.59 A	+0.85%	+7.25 A	+10.54%	1.1 ms
		500 V–300 V	+0.64 A	+0.93%	−4.75 A	−6.90%	1.2 ms
	1.65 kW	300 V–500 V	+0.57 A	+1.66%	+3.82 A	+11.11%	1.0 ms
		500 V–300 V	+0.62 A	+1.81%	−2.25 A	−6.54%	1.2 ms
	0.33 kW	300 V–500 V	+0.055 A	+0.8%	+1.85 A	+26.91%	1.33 ms
		500 V–300 V	+0.069 A	+1.0%	−1.87 A	−27.20%	1.40 ms

shows the transient analysis during simulation conditions of the load current deviations, %avg load ripples, transient dip, and %avg dip of the current during line regulation (i.e., 300 V_in–500 V_in and 500 V_in–300 V_in) tabulated by maintaining the load voltage at 48 V₀ and the load current at 68.75 A, 38.375 A, and 6.875 A, respectively. When the load was 3.3 kW, for (300 V_in–500 V_in) and (500 V_in–300 V_in), the max LCD was found to be +0.59 A and +0.64 A, and the %ALCR was +0.85% and +0.93%, respectively. The transient current dip was found to be +7.25 A, −4.75 A, and its average percentage dip was +10.54%, −6.90%, respectively, with a settling time of 1.1 ms and 1.2 ms. In the case of 1.65 kW, the max LCD was found to be +0.57 A and +0.62 A, and the %ALCR was +1.66% and +1.81%, respectively. The transient current dip was found to be +3.82 A, −2.25 A, and its average percentage dip was +11.11%, −6.54%, respectively, with a settling time of 1.0 ms and 1.2 ms. Lastly, in the case of 0.33 kW, the max LCD was found to be +0.055 A and +0.069 A, and the %ALCR was +0.8% and +1.0%, respectively. The transient current dip was found to be +1.85 A, −1.87 A, and its average percentage dip was +26.91%, −27.20%, respectively, with a settling time of 1.33 ms and 1.40 ms.

In addition, line regulations were conducted experimentally under variable input with a full load condition at 3.3 kW.

Table 3. Simulation vs. experimental analysis during line regulation.

Conditions	Max. LVD	%AVR	Max. LCD	%ACR
Simulations	+0.40 V	+0.83%	+0.59 A	+0.85%
Experimental	+0.76 V	+1.58%	+1.38 A	+2.00%

presents the comparative analysis of simulation and experimental conditions [24]. The simulation and experimental results indicate that the steady-state voltage deviation was +0.4 V (+0.83%) and +0.76 V (+1.58%), while the steady-state current deviation was +0.59 A (+0.85%) and +1.38 A (+2.0%), correspondingly.

The measurements in

Table 4. Simulation vs. experimental transient analysis during line regulation.

Conditions	Voltage Dip	Avg% of Dip	Current Dip	Avg% of Dip	Settling Time
Simulations	+5.20 V	+10.83%	+7.25 A	+10.54%	1.1 ms
Experimental	+6.17 V	+12.85%	+8.29 A	+12.05%	18 ms

indicate that there was a transient voltage dip (TVD) of +5.2 V (+10.83%), +6.17 V (+12.85%) and a transient current dip (TCD) of +7.25 A (+10.54%), +8.29 A (+12.05%), which are the transient voltage ripple (TVR) percentage and transient current ripple (TCR) percentage, respectively. The computational and experimental results indicate that the transient dip had a duration of 1.1 ms and 18 ms, respectively.

5.3. Discussion

The presented technology has the merit of scalability in operation, where it can operate in a wide range; ease of adaptability to any type of charging environment; and a high efficiency of operation. The major constraint of this technology is basically the construction of a complex system, and the management of the control algorithms for two bridges with maintaining the proper deadtime is the major challenge. The complexity of managing the two resonant tanks with hybrid structures will leads to a high cost of the system. The assembly, manufacturing, and fabrication of the proposed system having the higher involvement of complex strategies to minimize the losses during the development phase; this happens due to the system’s complexity.

6. Conclusions

This study focused on analyzing the iL²C resonant DC-DC converter, which utilizes a hybrid control method known as the variable frequency + phase shift modulation (VFPSM) technique. This research also contributes to the advancement of EV charging technology, ensuring efficient and reliable charging for xEVs. The converter is designed to accommodate a wide range of input voltages and is specifically intended for use in EV charging systems. This paper provides an explanation of the modeling and operational principles of the converter, and it conducts simulations of the whole proposed design using MATLAB/Simulink. The performance of the iL²C converter is evaluated under several conditions, and initially the converter is tested at various load and line conditions. Furthermore, the performance of the converter is verified through experimental validation utilizing GaN switches, and the results indicate that the controller effectively regulates the system. In future expansion of this work, predicative control techniques are to be implemented and analyzed.