**1. Introduction**

The research about plug-in and contactless charging of e-vehicles is increasing with the increase in the use of e-vehicles all over the world [1]. Depending on the power level and the usage area of e-vehicles, static or dynamic wireless charging can be used. In these power transmission systems, which are loosely coupled due to the height of the vehicle subchassis from the ground, the compensation is used on the primary side and secondary side to increase the power transmission efficiency. The main compensation topologies such as Series-Series, Series-Parallel (SP), and also the popular topologies that have additional components such as Inductor/Capacitor/Capacitor (LCC) [2], Inductor/Capacitor/Inductor (LCL) [3,4] etc. have been applied on the primary and secondary sides. The SS topology was preferred in this paper, since it is frequently chosen in the literature at similar power levels and uses fewer components. The other important components in the IPT structure are converter structures. In an IPT system supplied from a 50–60 Hz AC grid, the dualstage converters are traditionally preferred on the primary side (Figure 1). Particularly in recent years, there have been researchers who prefer the single-stage converter on the primary side [5,6]. By using the matrix converter on the IPT, a more compact primary side is provided independently of the lifetime and cost of the DC link capacitor. However, the dual-stage converter can be designed and controlled independently at each stage and also operates with high efficiency even at variable loads. Thus, this topology has been preferred in many industrial applications [7].

**Citation:** Yildiriz, E.; Bayraktar, M. Design and Implementation of a Wireless Charging System Connected to the AC Grid for an E-Bike. *Energies* **2022**, *15*, 4262. https://doi.org/ 10.3390/en15124262

Academic Editors: Adel El-Shahat and Byoung Kuk Lee

Received: 2 February 2022 Accepted: 4 April 2022 Published: 9 June 2022

**Publisher's Note:** MDPI stays neutral with regard to jurisdictional claims in published maps and institutional affiliations.

**Copyright:** © 2022 by the authors. Licensee MDPI, Basel, Switzerland. This article is an open access article distributed under the terms and conditions of the Creative Commons Attribution (CC BY) license (https:// creativecommons.org/licenses/by/ 4.0/).

change since changes.

**Figure 1.** Primary side configurations of IPT: (**a**) Dual‐stage; and (**b**) Single‐stage. **Figure 1.** Primary side configurations of IPT: (**a**) Dual-stage; and (**b**) Single-stage.

The charging current and voltage can be controlled from the primary side, the sec‐ ondary side, or both sides in case of misalignment or load changes. If the control is un‐ dertaken on the secondary side, the desired control can be achieved with the current and voltage data of the load. The control is achieved with an added DC/DC converter or us‐ ing a controlled rectifier on the secondary side [9,10]. That is, the control from the sec‐ ondary side does not require communication components but creates a more complex secondary side. When performing CC/CV control from the primary side, the variation of the load current and voltage must be known or estimated. These data can be transferred The battery characteristics should be taken into account when designing an IPT for the electrical device to be wirelessly charged. For the battery to operate in a healthy and long-lasting manner, it must be charged in Constant Current (CC) and Constant Voltage (CV) mode according to the battery charge profile [8]. During the charging process, the equivalent load resistance (*Ro*) is not constant as the charging current and voltage change. Although the mutual inductance between the coils (*M*) is constant in the full-alignment state, the transferred power, efficiency, charging current, and voltage also change since *R<sup>o</sup>* changes.

full‐alignment state, the transferred power, efficiency, charging current, and voltage also

from the secondary side to the primary side via a wireless communication link [11,12]. However, the data transmission security may be compromised in cases such as misa‐ lignment or the presence of any foreign objects. The parameter estimation approach can be based on the load estimation [13,14] and/or the mutual inductance estimation [15]. The CC/CV control from the primary side is generally based on operating the H‐bridge in‐ verter with variable frequency or the phase‐shift method [16,17]. The variable frequency operation may affect the power transfer performance and the efficiency due to diver‐ gence from the resonance frequency. The phase‐shifting method is the most widely used [4,18]. However, the phase‐shifting method does not allow the use of a standard soft‐switched inverter. Moreover, achieving the secondary side control with the phase‐shifting method on the primary side as in the e‐bike application focused on in this study (an IPT with a DC‐link voltage of 400 V on the primary side and also a low voltage on the secondary side) is very difficult. This is because the change sensitivity of the phase‐shift angle must be very high. Therefore, the sensitivity of the phase‐shifting limits the secondary side DC link voltage. Consequently, a DC/DC converter is required for the CC/CV control from the primary side of IPT applications, such as the wireless charging developed for the e‐bike and sourced from a 230 V AC grid. Nowadays, the research on wireless charging of e‐vehicles is generally focused on a power level of 3.3 kW and above. The primary side and secondary side DC‐link voltages are usually 400 V in IPT designs for these power levels [19–22]. Whereas the primary side DC‐link voltage is determined according to the AC grid, the secondary side DC voltage is related to the battery bank voltage. The power level to be transferred in the charging of e‐bikes and the voltage level of the battery group are low. In e‐bike applications en‐ countered in the literature, the system responses have been examined by connecting a DC The charging current and voltage can be controlled from the primary side, the secondary side, or both sides in case of misalignment or load changes. If the control is undertaken on the secondary side, the desired control can be achieved with the current and voltage data of the load. The control is achieved with an added DC/DC converter or using a controlled rectifier on the secondary side [9,10]. That is, the control from the secondary side does not require communication components but creates a more complex secondary side. When performing CC/CV control from the primary side, the variation of the load current and voltage must be known or estimated. These data can be transferred from the secondary side to the primary side via a wireless communication link [11,12]. However, the data transmission security may be compromised in cases such as misalignment or the presence of any foreign objects. The parameter estimation approach can be based on the load estimation [13,14] and/or the mutual inductance estimation [15]. The CC/CV control from the primary side is generally based on operating the H-bridge inverter with variable frequency or the phase-shift method [16,17]. The variable frequency operation may affect the power transfer performance and the efficiency due to divergence from the resonance frequency. The phase-shifting method is the most widely used [4,18]. However, the phase-shifting method does not allow the use of a standard soft-switched inverter. Moreover, achieving the secondary side control with the phase-shifting method on the primary side as in the e-bike application focused on in this study (an IPT with a DC-link voltage of 400 V on the primary side and also a low voltage on the secondary side) is very difficult. This is because the change sensitivity of the phase-shift angle must be very high. Therefore, the sensitivity of the phase-shifting limits the secondary side DC link voltage. Consequently, a DC/DC converter is required for the CC/CV control from the primary side of IPT applications, such as the wireless charging developed for the e-bike and sourced from a 230 V AC grid.

supply to the inverter input [23–29], or a DC‐link voltage in the range of 60–85 V can be achieved on the primary side using a grid‐connected step‐down transformer [30]. In Nowadays, the research on wireless charging of e-vehicles is generally focused on a power level of 3.3 kW and above. The primary side and secondary side DC-link voltages are usually 400 V in IPT designs for these power levels [19–22]. Whereas the primary side DC-link voltage is determined according to the AC grid, the secondary side DC voltage is related to the battery bank voltage. The power level to be transferred in the charging of ebikes and the voltage level of the battery group are low. In e-bike applications encountered in the literature, the system responses have been examined by connecting a DC supply to the inverter input [23–29], or a DC-link voltage in the range of 60–85 V can be achieved on the primary side using a grid-connected step-down transformer [30]. In these papers, the CC/CV control was carried out with the phase-shifting method over the h-bridge inverter. Using a DC/DC converter instead of a step-down transformer at the inverter input provides a more compact primary pad. A dual-stage converter was used in [31], and a buck converter was preferred at the inverter input. Due to the operation limits of the buck converter, this wireless charging system requires the use of a DC/DC converter on the secondary side for the charge control. In this case, both the use of more components has emerged and the idea of a simple secondary side acquisition has been moved away from. Considering the researches on wireless charging of e-bikes that use low secondary side DC-link voltage, the main contributions of this paper are summarized as follows. these papers, the CC/CV control was carried out with the phase‐shifting method over the h‐bridge inverter. Using a DC/DC converter instead of a step‐down transformer at the inverter input provides a more compact primary pad. A dual‐stage converter was used in [31], and a buck converter was preferred at the inverter input. Due to the operation limits of the buck converter, this wireless charging system requires the use of a DC/DC con‐ verter on the secondary side for the charge control. In this case, both the use of more components has emerged and the idea of a simple secondary side acquisition has been moved away from. Considering the researches on wireless charging of e‐bikes that use low secondary side DC‐link voltage, the main contributions of this paper are summa‐ rized as follows. (1) A wireless charging system connected to the AC grid has been designed to meet


The organization of this paper is as follows. The optimum IPT design was carried out considering the charging requirements of the e-bike, as described in Section 2. Then, the design of the forward converter, in which the charge control and the no-load condition test were carried out, is presented. The simulation and experimental results are presented in Section 4, comparatively. Finally, Section 5 concludes this paper. the design of the forward converter, in which the charge control and the no‐load condi‐ tion test were carried out, is presented. The simulation and experimental results are presented in Section 4, comparatively. Finally, Section 5 concludes this paper. **2. Design of Inductive Power Transfer System for E‐Bikes**

#### **2. Design of Inductive Power Transfer System for E-Bikes** The limits of electrical parameters such as DC‐link voltages and resonance fre‐

*Energies* **2022**, *15*, x FOR PEER REVIEW 3 of 16

The limits of electrical parameters such as DC-link voltages and resonance frequency, as well as physical criteria such as the air-gap of the power transfer, should be determined correctly when starting an IPT system design. Wireless charging of a 250 W e-bike with 36 V, 20 Ah gel batteries was investigated in this paper. The charging system's input is connected to a 230 V AC grid. The general scheme of the system is given in Figure 2. According to the battery characteristics, the maximum voltage needed to charge the 3 × 12 V battery pack is 44 V. When charging at a constant current, the charging current is required to be 2.5 A. quency, as well as physical criteria such as the air‐gap of the power transfer, should be determined correctly when starting an IPT system design. Wireless charging of a 250 W e‐bike with 36 V, 20 Ah gel batteries was investigated in this paper. The charging sys‐ tem's input is connected to a 230 V AC grid. The general scheme of the system is given in Figure 2. According to the battery characteristics, the maximum voltage needed to charge the 3 × 12 V battery pack is 44 V. When charging at a constant current, the charging cur‐ rent is required to be 2.5 A.

**Figure 2.** Structural diagram of the IPT proposed for e‐bike wireless charging. **Figure 2.** Structural diagram of the IPT proposed for e-bike wireless charging.

The secondary side output voltage of the IPT, ′ is calculated using Equation (1) according to the battery charge requirements. The primary and secondary side voltages The secondary side output voltage of the IPT, *V<sup>o</sup>* 0 is calculated using Equation (1) according to the battery charge requirements. The primary and secondary side voltages are

are expected to be close to each other for optimum coil usage [32]. The primary side voltage was selected considering this closeness relation. There is a relationship be‐

expected to be close to each other for optimum coil usage [32]. The primary side voltage *V<sup>P</sup>* was selected considering this closeness relation. There is a relationship between the output voltage of the DC-DC converter (*VF*) and *V<sup>P</sup>* as in Equation (2). Accordingly, a forward converter was preferred in this study in order to reduce the output voltage of the rectifier connected to the AC grid to the desired DC voltage level.

$$V\_o' = \frac{\pi}{2\sqrt{2}} V\_o \tag{1}$$

$$V\_P = \frac{4}{\pi\sqrt{2}}V\_F\tag{2}$$

The voltage equations of the primary and secondary side are written as Equations (3) and (4) for SS topology. Here *ω* is the resonance frequency. The self-inductances (*L<sup>P</sup>* and *LS*) of the primary and secondary side windings depend on the winding dimensions and the number of turns, which are decided according to the coil design. Primary and secondary resonance capacitors (*C<sup>P</sup>* and *CS*) are determined considering *LP*, *LS*, and *ω*. Another important parameter in voltage equations, *Ro*, is the equivalent resistance at the rectifier input and is calculated from Equation (5). The primary and secondary sides are expected to operate in resonance during the power transmission. Thus, the efficiency of the power transfer can be calculated from (6). Using Equation (4), the ratio between primary and secondary side currents is written as in (7). Equation (8) is obtained when this ratio is used in Equation (6). Consequently, the change in load and mutual inductance directly affects the power transfer efficiency (PTE).

$$V\_P = R\_P I\_P + j\left(L\_P \omega - \frac{1}{\mathbb{C}\_P \omega}\right) I\_P + j\omega M I\_S \tag{3}$$

$$j\omega MI\_P = R\_S I\_S + j\left(L\_S \omega - \frac{1}{C\_S \omega}\right) I\_S + R\_o I\_S \tag{4}$$

$$R\_o = \frac{8}{\pi^2} R\_L \tag{5}$$

$$PTE = \frac{R\_o I\_S^2}{R\_P I\_P^2 + R\_S I\_S^2 + R\_o I\_S^2} \tag{6}$$

$$\frac{I\_P}{I\_S} = \frac{R\_S + R\_o}{\omega M} \tag{7}$$

$$PTE = \frac{R\_o}{R\_P \left(\frac{I\_P}{I\_S}\right)^2 + (R\_S + R\_o)} = \frac{R\_o}{R\_P \left(\frac{R\_S + R\_o}{\omega M}\right)^2 + (R\_S + R\_o)}\tag{8}$$

The resonance frequency and mutual inductance should be high for maximum PTE, as seen in Equation (9). However, as the operating frequency increases, the effective series resistance (ESR) of the coils increases too. Therefore, quality factors should also be taken into account in the PTE calculation. The quality factors of the primary and secondary sides must be carefully selected to avoid bifurcation during charging. The quality factors are calculated as in Equation (10) for the SS topology. *Q<sup>P</sup>* and *Q<sup>S</sup>* values are usually chosen to be close to each other and to be *Q<sup>P</sup>* > *Q<sup>S</sup>* [33,34].

$$
\omega M \gg \sqrt{R\_P} (R\_S + R\_O) \tag{9}
$$

$$Q\_P = \frac{L\_P R\_O}{\omega M^2}, \ Q\_S = \frac{wL\_S}{R\_O} \tag{10}$$

In the optimum IPT design, besides electrical constraints such as input and output voltages and required power, the physical constraints should also be taken into account. The maximum outer dimensions of the coils and power transmission height are known,

due to the sub-chassis size limitation. In order to protect the secondary winding fixed to the sub-chassis of the e-bike, a plastic protective cover according to the winding dimensions was prepared. The height of the impact-proof plastic cover was 10 cm from the ground. The maximum area that the secondary winding could use was 240 × 280 mm, considering the sub-chassis limits. In order to make the most of this area, a rectangular winding pair was preferred. In this study, a coil pair with an air core was chosen in order to avoid additional weight on the bicycle. Thus, since the winding inductances can be calculated analytically, not FEA, the optimum winding pair could be determined quickly.

In addition to physical constraints, design constraints such as winding current densities, maximum frequency, and the avoidance of bifurcation should also be considered. It is possible to use smaller winding pairs for the same power transfer as the operating frequency increases. However, the magnetic flux density outside the power transmission region also increases with the operating frequency. The maximum operating frequency of the optimal IPT to be designed was selected at around 85 kHz, considering compliance with ICNIRP and IEEE c95.1 standards and SAE J2954 criteria. The maximum winding current density was determined as 3 A/mm<sup>2</sup> in order to keep the thermal effect small.

The winding designs capable of transmitting the desired power were scanned with an algorithm as in [33], considering the determined design constraints and physical constraints. The self and mutual inductances of the air-core windings were calculated analytically. The winding inductances depend on the winding dimensions and the number of turns. The *K<sup>D</sup>* parameter was used to determine the best winding pair among the winding pairs that could transfer the desired power. The *K<sup>D</sup>* parameter is a design factor, which is determined by the winding quality factors and the maximum operating frequency, and defined as the winding utilization factor [35].

The parameters of the optimum IPT are given in Table 1. The windings were wound with 38 AWG litz wire, taking into account the operating frequency and the current density. When a load close to the equivalent load was connected to the secondary side, the current and voltages close to the design values were measured experimentally. In the experimental study, the winding resistances were high due to the additional connection lengths. This situation was also reflected in the observed experimental efficiency.


**Table 1.** Design parameters of the optimum IPT.

The load in the designed IPT system is not static. The equivalent resistance value will also change with the battery charge level. The current and voltage gains for different loads were calculated according to the design parameters. Due to the nature of the SS topology, an uncontrolled IPT system operates in CC mode. Since the magnitudes of the currentvoltage gain *GI*−*<sup>V</sup>* are small, a limited change in current is observed in the load changes that may occur while operating at the resonance frequency (as seen in Figure 3a). When the voltage gain for load changes is examined, it is seen that *GV*−*<sup>V</sup>* increases significantly as the resistance value increases at 85 kHz, for which the IPT is designed (Figure 3b). In

0

0.05

0.05

0.1

0.1

0.15

0.15

Gain | Gi-v |

Gain | Gi-v |

0.2

0.2

0.25

0.25

voltage can reach dangerous points. 5 Ohm 20 Ohm 40 Ohm 80 Ohm 1 2 3 4 5 Gain | Gv-v | 5 Ohm 20 Ohm 40 Ohm 80 Ohm 60 65 70 75 80 85 90 95 100 105 110 5 Ohm 20 Ohm 40 Ohm 80 Ohm 60 65 70 75 80 85 90 95 100 105 110 0 1 2 3 4 5 Gain | Gv-v | 5 Ohm 20 Ohm 40 Ohm 80 Ohm

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other words, due to the nature of the SS topology, it cannot give a constant voltage output at variable loads. Therefore, especially when the secondary side is open-circuit, the output ure 3b). In other words, due to the nature of the SS topology, it cannot give a constant voltage output at variable loads. Therefore, especially when the secondary side is open‐circuit, the output voltage can reach dangerous points. voltage output at variable loads. Therefore, especially when the secondary side is open‐circuit, the output voltage can reach dangerous points.

nificantly as the resistance value increases at 85 kHz, for which the IPT is designed (Fig‐

nificantly as the resistance value increases at 85 kHz, for which the IPT is designed (Fig‐ ure 3b). In other words, due to the nature of the SS topology, it cannot give a constant

0

(**a**) (**b**)

(**a**) (**b**)

#### **3. Forward Converter Design 3. Forward Converter Design 3. Forward Converter Design**

60 65 70 75 80 85 90 95 100 105 110

Frequency (kHz)

Frequency (kHz)

The forward converter isolates the input and output sides of the converter from each other using the forward transformer. 300 V DC voltages can be stepped down to 5–10 V using a forward converter. Considering the switching times of the semiconductor circuit elements, it is difficult to obtain such a duty cycle with a buck converter. Although the power levels of forward converters are limited, a forward converter can be used in cases in which the required power is low (44 V × 2.5 A), as in this study. The forward converter isolates the input and output sides of the converter from each other using the forward transformer. 300 V DC voltages can be stepped down to 5–10 V using a forward converter. Considering the switching times of the semiconductor circuit elements, it is difficult to obtain such a duty cycle with a buck converter. Although the power levels of forward converters are limited, a forward converter can be used in cases in which the required power is low (44 V × 2.5 A), as in this study. The forward converter isolates the input and output sides of the converter from each other using the forward transformer. 300 V DC voltages can be stepped down to 5–10 V using a forward converter. Considering the switching times of the semiconductor circuit elements, it is difficult to obtain such a duty cycle with a buck converter. Although the power levels of forward converters are limited, a forward converter can be used in cases in which the required power is low (44 V × 2.5 A), as in this study.

60 65 70 75 80 85 90 95 100 105 110

Frequency (kHz)

Frequency (kHz)

In the two‐switch forward converter, the current and voltage stress that the switches are exposed to is less than that of a single‐switch one. The equivalent circuit of the two‐switch forward converter according to the ON and OFF operating modes is shown in Figure 4. When the switches (Q1 and Q2) are ON, energy is transferred depending on the conver‐ sion ratio from primary to secondary of the transformer. At this moment, D3 is ON, whereas D4 is OFF. When the switches turn off, the primary winding of the transformer is reversely connected to the input voltage via D1 and D2. Thus, there is no need to use an additional winding to reset the transformer. On the secondary side, the voltage of ி reverses. Therefore, D3 turns off and D4 turns on. Thus, the linearly decreasing current of ி continues to flow. The output voltage of the forward converter is calculated from Equation (11). Here ி, ி, denote the converter efficiency, forward‐turn ratio and duty ratio, respectively. The turn ratio of the transformer is calculated taking into account the maximum of the duty ratio and the minimum input voltage. ி and ி are calcu‐ lated according to the limited current and voltage ripple. In the two-switch forward converter, the current and voltage stress that the switches are exposed to is less than that of a single-switch one. The equivalent circuit of the twoswitch forward converter according to the ON and OFF operating modes is shown in Figure 4. When the switches (Q1 and Q2) are ON, energy is transferred depending on the conversion ratio from primary to secondary of the transformer. At this moment, D3 is ON, whereas D4 is OFF. When the switches turn off, the primary winding of the transformer is reversely connected to the input voltage via D1 and D2. Thus, there is no need to use an additional winding to reset the transformer. On the secondary side, the voltage of *L<sup>F</sup>* reverses. Therefore, D3 turns off and D4 turns on. Thus, the linearly decreasing current of *L<sup>F</sup>* continues to flow. The output voltage of the forward converter is calculated from Equation (11). Here *ηF*, *NF*, *D* denote the converter efficiency, forward-turn ratio and duty ratio, respectively. The turn ratio of the transformer is calculated taking into account the maximum of the duty ratio and the minimum input voltage. *L<sup>F</sup>* and *C<sup>F</sup>* are calculated according to the limited current and voltage ripple. In the two‐switch forward converter, the current and voltage stress that the switches are exposed to is less than that of a single‐switch one. The equivalent circuit of the two‐switch forward converter according to the ON and OFF operating modes is shown in Figure 4. When the switches (Q1 and Q2) are ON, energy is transferred depending on the conver‐ sion ratio from primary to secondary of the transformer. At this moment, D3 is ON, whereas D4 is OFF. When the switches turn off, the primary winding of the transformer is reversely connected to the input voltage via D1 and D2. Thus, there is no need to use an additional winding to reset the transformer. On the secondary side, the voltage of ி reverses. Therefore, D3 turns off and D4 turns on. Thus, the linearly decreasing current of ி continues to flow. The output voltage of the forward converter is calculated from Equation (11). Here ி, ி, denote the converter efficiency, forward‐turn ratio and duty ratio, respectively. The turn ratio of the transformer is calculated taking into account the maximum of the duty ratio and the minimum input voltage. ி and ி are calcu‐ lated according to the limited current and voltage ripple.

**Figure 4.** The equivalent circuits of the two‐switch forward converter: (**a**) ON state; (**b**) OFF state. **Figure 4.** The equivalent circuits of the two‐switch forward converter: (**a**) ON state; (**b**) OFF state. **Figure 4.** The equivalent circuits of the two-switch forward converter: (**a**) ON state; (**b**) OFF state.

The voltage ripple of the converter output will further increase the voltage ripple in the charging process. Therefore, the magnitude of the capacitor at the converter output is important (12). Here *ω<sup>F</sup>* is the forward switching frequency, and ∆*I<sup>F</sup>* and ∆*V<sup>F</sup>* represent the ripple in current and voltage of the forward converter output, respectively. The input of the forward converter is connected to a single-phase full-wave rectifier. The parameters of the designed forward converter are given in Table 2.

$$V\_F = \eta D V\_{IN} N\_F \tag{11}$$

$$\mathcal{C}\_{\text{F}} = \frac{\Delta I\_{\text{F}}}{\omega\_{\text{F}} \Delta V\_{\text{F}}} \tag{12}$$

**Table 2.** The parameters of the forward converter.

