*3.4. Analysis of Efficiency*

The efficiencies of switching losses, conduction losses, and split-inductors losses are considered. The switching energies obtained from the DPT test are used for switching loss calculation. In case of switching loss, turn ON loss *Pon* and turn OFF loss *Poff* are determined by:

$$P\_{\rm on} = f\_{\rm s} \times E\_{\rm on} \tag{12}$$

$$P\_{off} = f\_{s} \times E\_{off} \tag{13}$$

where the switching energies *Eon* and *Eoff* can be determined by:

$$E\_{\rm on} = I\_{\rm s} \times \mathbf{x}\_{\rm on} \tag{14}$$

$$E\_{off} = I\_s \times \mathbf{x}\_{off} \tag{15}$$

The equations for *xon* and *xoff* can be mathematically expressed by:

*xon* = *x*1*on* × *I<sup>s</sup>* <sup>2</sup> <sup>+</sup> *<sup>x</sup>*2*on* <sup>×</sup> *<sup>I</sup><sup>s</sup>* <sup>+</sup> *<sup>x</sup>*3*on* (16) *xo f f* = *x*1*o f f* × *I<sup>s</sup>* <sup>2</sup> <sup>+</sup> *<sup>x</sup>*2*o f f* <sup>×</sup> *<sup>I</sup><sup>s</sup>* <sup>+</sup> *<sup>x</sup>*3*o f f* (17) *Micromachines* **2021**, *12*, x FOR PEER REVIEW 14 of 20 (**a**) (**b**) (**c**) (**d**)

 $\textbf{Figure 11.}$ 
 $\text{Switching curves of } \\$\_2 \text{ obtained from LTSpace: (a) switch turn ON for conventional ANPC, (b) switch turn OFF: (c) switch turn ON for hybrid ANPC, (d) switch turn OFF for hybrid ANPC.}$ 

ANPC inverter. This has resulted in higher turn OFF losses incurred by the conventional inverter. Although the margin of differences between the conventional inverter and the hybrid ANPC for turn OFF losses is very close, the overall switching losses of hybrid ANPCs are significantly lower because the turn ON losses are more dominant. *3.4. Analysis of Efficiency*  Here, the constants *x*1*on*, *x*2*on*, *x*3*on*, . . . . . . are representative of the constants that are used for curve fitting shown in Figure 10. Additionally, the conduction losses are calculated using the manufacturer's datasheet curves for different load currents in the case of conventional Si switches, whereas, for Ga2O<sup>3</sup> switches, it has been obtained from the information provided in [35]. The expressions obtained from these curves are:

$$P\_{\mathfrak{c}} = \mathfrak{x}\_{4} \times I\_{\mathfrak{s}}^{2} + \mathfrak{x}\_{5} \times I\_{\mathfrak{s}} \tag{18}$$

<sup>ଶ</sup> + ݔଶ × ܫ௦ + ݔଷ (16)

<sup>ଶ</sup> + ݔଶ × ܫ௦ + ݔଷ (17)

calculation. In case of switching loss, turn ON loss *Pon* and turn OFF loss *Poff* are determined by: (12) ܧ × ௦݂ = ܲ where *x*<sup>4</sup> and *x*<sup>5</sup> are the constants for the curve fitting of Figure 10. Furthermore, the core losses from the split inductors are determined using the curves shown in Figure 8 and the information provided in [34].

considered. The switching energies obtained from the DPT test are used for switching loss

 (13) ܧ × ௦݂ = ܲ where the switching energies *Eon* and *Eoff* can be determined by: (14) ݔ × ௦ܫ = ܧ (15) ݔ × ௦ܫ = ܧ In this paper, the losses of both conventional ANPCs as well as the proposed hybrid ANPC inverter are calculated considering different loads. In addition, three switching frequencies are considered to compare the loss behavior of the configurations, as shown in Figure 12. It can be validated from Figure 12 that because of using UWBG switches and due to reduced switching losses, the proposed inverter's efficiency in all cases is much higher compared to the conventional Si-based ANPC inverter.

The equations for *xon* and *xoff* can be mathematically expressed by: ௦ܫ × ଵݔ = ݔ

higher compared to the conventional Si-based ANPC inverter.

mation provided in [35]. The expressions obtained from these curves are:

information provided in [34].

௦ܫ × ସݔ = ܲ

where *x*4 and *x*5 are the constants for the curve fitting of Figure 10. Furthermore, the core losses from the split inductors are determined using the curves shown in Figure 8 and the

In this paper, the losses of both conventional ANPCs as well as the proposed hybrid ANPC inverter are calculated considering different loads. In addition, three switching frequencies are considered to compare the loss behavior of the configurations, as shown in Figure 12. It can be validated from Figure 12 that because of using UWBG switches and due to reduced switching losses, the proposed inverter's efficiency in all cases is much

**Figure 12.** Efficiency comparison between conventional ANPC and hybrid ANPC under various switching frequencies. **Figure 12.** Efficiency comparison between conventional ANPC and hybrid ANPC under various switching frequencies.

Here, the constants *x1on*, *x2on*, *x3on*, ....... are representative of the constants that are used for curve fitting shown in Figure 10. Additionally, the conduction losses are calculated using the manufacturer's datasheet curves for different load currents in the case of conventional Si switches, whereas, for Ga2O3 switches, it has been obtained from the infor-

<sup>ଶ</sup> + ݔହ × ܫ௦ (18)

### *3.5. Analysis of High-Frequency Transient in Output Voltage 3.5. Analysis of High-Frequency Transient in Output Voltage*

Along with the advantages of the conventional ANPC inverter, the proposed inverter can reduce high-frequency switching noise in the output voltage. This high-frequency noise primarily contributes to electromagnetic interference (EMI) issues and also has some impacts on the operation of the gate driver [24]. In addition, the incorporation of the two split inductors, i.e., *L*1 and *L*2, in the proposed inverter topology makes it possible to decrease the high-frequency transients considerably because of the filter of the transients by the inductances. Thus, the size of the electromagnetic compatibility (EMC) filter becomes significantly smaller. This statement can be validated by using (1) and (2). If any sudden change has occurred in the output voltage of the presented inverter, that impact will be damped by the inductance's inherent capability to oppose any sudden change in current. The blocking voltage is tuned according to the values of the split inductor. For the proposed design, as the inductance value was 1 uH for the split inductor, the output voltages' harmonic spectra can be illustrated for both conventional ANPCs and the proposed hybrid ANPC inverter through LT Spice simulation, as is illustrated in Figure 13. It can be seen that the final range of the high-frequency transient will be 5 to 15 MHz. This is due to the damped high-frequency voltage in this frequency range by the split-inductors. Along with the advantages of the conventional ANPC inverter, the proposed inverter can reduce high-frequency switching noise in the output voltage. This high-frequency noise primarily contributes to electromagnetic interference (EMI) issues and also has some impacts on the operation of the gate driver [24]. In addition, the incorporation of the two split inductors, i.e., *L*<sup>1</sup> and *L*2, in the proposed inverter topology makes it possible to decrease the high-frequency transients considerably because of the filter of the transients by the inductances. Thus, the size of the electromagnetic compatibility (EMC) filter becomes significantly smaller. This statement can be validated by using (1) and (2). If any sudden change has occurred in the output voltage of the presented inverter, that impact will be damped by the inductance's inherent capability to oppose any sudden change in current. The blocking voltage is tuned according to the values of the split inductor. For the proposed design, as the inductance value was 1 uH for the split inductor, the output voltages' harmonic spectra can be illustrated for both conventional ANPCs and the proposed hybrid ANPC inverter through LT Spice simulation, as is illustrated in Figure 13. It can be seen that the final range of the high-frequency transient will be 5 to 15 MHz. This is due to the damped high-frequency voltage in this frequency range by the split-inductors. Thus, the added split inductors for the shoot-through protection also help to reduce the EMI filter size. *Micromachines* **2021**, *12*, x FOR PEER REVIEW 16 of 20 Thus, the added split inductors for the shoot-through protection also help to reduce the EMI filter size.

**Figure 13.** High-frequency voltage spectrum of the conventional ANPC and hybrid ANPC. **Figure 13.** High-frequency voltage spectrum of the conventional ANPC and hybrid ANPC.

1.2566 × 10ି × 6000 × 400 = 2.08 × 10ିmଶ (20)

The cross-sectional area (*A*) of an EMI Filter for the hybrid ANPC with 100 kHz

Here, *r*, *μ0*, *μr*, and *N* represent the toroid radius to centerline, the magnetic constant, the relative permeability of Mn–Zn ferrite, and the number of turns, respectively. Similarly, the cross-sectional area of the EMI filter for a conventional ANPC can be calculated

Thus, it can be observed that the size of the EMI filter for the proposed ANPC inverters becomes halved compared to the conventional ANPC inverter due to the usage of split inductors. Furthermore, the relative permeability versus the switching frequency curve for Mn–Zn ferrite is shown in Figure 14. It is noticeable that with higher switching frequency, the relative permeability tends to decrease logarithmically. Therefore, the crosssectional area of the EMI filter will increase with a higher switching frequency.

The MATLAB/Simulink version of the proposed hybrid ANPC inverter is developed in this section to validate that it can operate in various ranges of power factors. LTSpice simulation is not required in this case since this feature is embraced by the proposed inverter due to implementing the split-inductors-based design, and this feature is not associated with using UWBG switches. Thus, for operational simplicity, MATLAB Simulink along with ideal MOSFETs and IGBTs are used to develop the proposed inverter. The output voltage and current waveforms are obtained for the proposed topology using a 200 V DC link. Thus, a voltage of 100 V will come across each DC-link capacitor. The simulation tests are repeated with the loads with non-unity power factor. To show the applicability of the proposed converter compared to the existing topologies. The results show the non-distorted waveforms for voltage and currents. The results for output voltage *Van* and load current *I*a are shown in Figure 15. Furthermore, the voltage across one DC-link capacitor is also shown, which indicates the nature of the common-mode voltage

ଽ10ି × 50 × 20 × 0.1 × ߨ2

switching frequency can be determined by: ܮݎߨ2

ܮݎߨ2

 =ଶܰߤߤ

 =ଶܰߤߤ

*3.6. Analysis of Operation at Various Range of Power Factors* 

= ܣ

= ܣ

as follows:

The cross-sectional area (*A*) of an EMI Filter for the hybrid ANPC with 100 kHz switching frequency can be determined by:

$$A = \frac{2\pi rL}{\mu\_0 \mu\_r N^2} = \frac{2\pi \times 0.1 \times 0.5 \times 10^{-6}}{1.2566 \times 10^{-6} \times 6000 \times 400} = 1.04 \times 10^{-7} \text{m}^2 \tag{19}$$

Here, *r*, *µ0*, *µ<sup>r</sup>* , and *N* represent the toroid radius to centerline, the magnetic constant, the relative permeability of Mn–Zn ferrite, and the number of turns, respectively. Similarly, the cross-sectional area of the EMI filter for a conventional ANPC can be calculated as follows:

$$A = \frac{2\pi rL}{\mu\_0 \mu\_r N^2} = \frac{2\pi \times 0.1 \times 20 \times 50 \times 10^{-9}}{1.2566 \times 10^{-6} \times 6000 \times 400} = 2.08 \times 10^{-7} \text{m}^2\tag{20}$$

Thus, it can be observed that the size of the EMI filter for the proposed ANPC inverters becomes halved compared to the conventional ANPC inverter due to the usage of split inductors. Furthermore, the relative permeability versus the switching frequency curve for Mn–Zn ferrite is shown in Figure 14. It is noticeable that with higher switching frequency, the relative permeability tends to decrease logarithmically. Therefore, the cross-sectional area of the EMI filter will increase with a higher switching frequency. *Micromachines* **2021**, *12*, x FOR PEER REVIEW 17 of 20 (CMV). It can be observed that the CMV is always constant at 100 V and it does not contain any ripples of high frequency. Thus, the leakage-current-related issues can also be solved using this topology.

**Figure 14.** Relative permeability of Mn Zn ferrite under different switching frequencies. **Figure 14.** Relative permeability of Mn Zn ferrite under different switching frequencies.

*3.6. Analysis of Operation at Various Range of Power Factors*

The MATLAB/Simulink version of the proposed hybrid ANPC inverter is developed in this section to validate that it can operate in various ranges of power factors. LTSpice simulation is not required in this case since this feature is embraced by the proposed inverter due to implementing the split-inductors-based design, and this feature is not associated with using UWBG switches. Thus, for operational simplicity, MATLAB Simulink along with ideal MOSFETs and IGBTs are used to develop the proposed inverter. The output voltage and current waveforms are obtained for the proposed topology using a 200 V DC link. Thus, a voltage of 100 V will come across each DC-link capacitor. The simulation tests are repeated with the loads with non-unity power factor. To show the applicability of the proposed converter compared to the existing topologies. The results show the non-distorted waveforms for voltage and currents. The results for output voltage *Van* and load current *Ian* are shown in Figure 15. Furthermore, the voltage across one DC-link capacitor is also shown, which indicates the nature of the common-mode voltage (CMV). It can be observed that the CMV is always constant at 100 V and it does not contain any ripples of high frequency. Thus, the leakage-current-related issues can also be solved using this topology.

(**a**) (**b**)

**Figure 15.** Simulation results for the hybrid ANPC inverter for the output voltage (Van), current (Ian) and common-mode voltage (CMV) with (**a**) unity power factor, (**b**) non-unity power factor.

To sum up, this paper presents a three-level hybrid ANPC topology that includes Ga2O3-based MOSFET as well as Si-based IGBTs. This inverter has split inductors at the

**4. Conclusions** 

**Figure 14.** Relative permeability of Mn Zn ferrite under different switching frequencies.

10 100 1000 10000

Switching frequency (kHz)

(CMV). It can be observed that the CMV is always constant at 100 V and it does not contain any ripples of high frequency. Thus, the leakage-current-related issues can also be solved

**Figure 15.** Simulation results for the hybrid ANPC inverter for the output voltage (Van), current (Ian) and common-mode voltage (CMV) with (**a**) unity power factor, (**b**) non-unity power factor. **Figure 15.** Simulation results for the hybrid ANPC inverter for the output voltage (*Van*), current (*Ian*) and common-mode voltage (CMV) with (**a**) unity power factor, (**b**) non-unity power factor.

### **4. Conclusions 4. Conclusions**

using this topology.

100

1000

Relative Permiability

10000

To sum up, this paper presents a three-level hybrid ANPC topology that includes Ga2O3-based MOSFET as well as Si-based IGBTs. This inverter has split inductors at the To sum up, this paper presents a three-level hybrid ANPC topology that includes Ga2O3-based MOSFET as well as Si-based IGBTs. This inverter has split inductors at the output, which are not only capable of protecting against the shoot-through fault but can also contribute to the reduced EMI in the output voltage. To maximize the efficiency of our converter, as well as to maximize the benefit of the Ga2O<sup>3</sup> switches, both the modulation technique as well as four modes of operation are discussed in this paper. The efficiency of both the conventional ANPC and the proposed hybrid ANPC inverter is measured and compared through LT Spice and MATLAB simulations. It was observed that under various switching frequencies and output power, the minimum efficiency was 96.8%, whereas a 99.1% maximum efficiency was obtained by the proposed inverter. The employability of the proposed module is analyzed by taking into consideration the reduced overshoots in switching waveforms, higher efficiency, lower current, voltage stress, minimized shootthrough current, and EMI. Eliminating the dominating switching losses, especially turn-on losses, as well as the addition of UWBG switches, contributes to an increase in efficiency. In addition, to validate the inverter's capability to supply reactive power, the module was operated under both various load conditions by changing the power factors. The simulation result acquired from the proposed module coincides with the theoretical results. The following is a list of the manuscript's concluding statements:


This study applies UWBG switches for ANPC inverters considering the technical pros and cons. Since the fabrication and production of UWBG semiconductors are still in their early phase industrially, experimental verification of the proposed inverter will be considered in the future. In the future, UWBG devices have great potential in the field of power electronics because of their superior characteristics over wide bandgap (WBG) and conventional semiconductors. Thus, researchers can utilize this opportunity to incorporate UWBG devices in other inverters/converter topologies and power electronic applications.

**Author Contributions:** Conceptualization, S.T.M. and J.I.; methodology, S.T.M. and N.Z.Y.; formal analysis, S.T.M. and M.S.H.L.; investigation, S.T.M., N.Z.Y. and M.S.H.L.; resources, N.Z.Y., J.I. and L.K.H.; data curation, S.T.M., L.K.H. and K.H.; writing—original draft preparation, S.T.M.; writing review and editing, M.S.H.L., K.H. and M.S.M.; S.A. and A.H.; supervision, N.Z.Y. and M.S.H.L.; project administration, M.S.H.L.; funding acquisition, M.S.H.L. All authors have read and agreed to the published version of the manuscript.

**Funding:** This work was supported by Universiti Kebangsaan Malaysia under Grant Code GP-2021-K023221.

**Conflicts of Interest:** The authors declare no conflict of interest.
