A High-Frequency Isolation (HFI) Charging DC Port Combining a Front-End Three-Level Converter with a Back-End LLC Resonant Converter

Abstract

The high-frequency isolation (HFI) charging DC port can serve as the interface between unipolar/bipolar DC buses and electric vehicles (EVs) through the two-power-stage system structure that combines the front-end three-level converter with the back-end logical link control (LLC) resonant converter. The DC output voltage can be maintained within the desired voltage range by the front-end converter. The electrical isolation can be realized by the back-end LLC converter, which has the bus converter function. According to the three-level topology, the low-voltage rating power devices can be adapted for half-voltage stress of the total DC grid, and the PWM phase-shift control can double the equivalent switching frequency to greatly reduce the filter volume. LLC resonant converters have advance characteristics of inverter-side zero-voltage-switching (ZVS) and rectifier-side zero-current switching (ZCS). In particular, it can achieve better performance under quasi-resonant frequency mode. Additionally, the magnetizing current can be modified following different DC output voltages, which have the self-adaptation ZVS condition for decreasing the circulating current. Here, the principles of the proposed topology are analyzed in detail, and the design conditions of the three-level output filter and high-frequency isolation transformer are explored. Finally, a 20 kW prototype with the 760 V input and 200–500 V output are designed and tested. The experimental results are demonstrated to verify the validity and performance of this charging DC port system structure.

1. Introduction

As solutions of energy crisis and environmental pollution, EVs that can run on alternative resources of energy have increasingly attracted attention for investigations of decreasing fossil fuel consumption and reducing greenhouse gas emissions [1,2]. The commercial success of EVs relies heavily on the presence of high-efficiency charging stations to increase mileage and shorten charging time [3,4,5]. Electric vehicle technologies involved with hybrid electric vehicles (HEVs), plug-in hybrid electric vehicles (PHEVs), and plug-in pure electric vehicles (PPEVs), such as the Toyota Prius and Lexus RX 400h, have been commercialized and are available in the market. A large number of high-power charging stations need to be constructed to solve consumers’ need for long-distance transport with electric vehicles (EVs). The construction of these facilities is a key factor in attracting more consumers to the use of EVs [6,7,8,9,10].

Recently, a distributed generation system, including the solar photovoltaic and wind power generation system as green energy, is playing an increasingly important role in energy structures. The combination of renewable energy sources with EV charging stations and energy storage systems is inevitable [11,12,13,14,15]. A high-efficiency, high-power-density, high-reliability, and cost-effective charging station, depicted in Figure 1, has been designed [16,17,18]. Additionally, a power electronics transformer (PET) can serve as the interface between medium- and low-voltage levels [19].

Chargers can be classified by two types—the on-board charger, located on the vehicle itself, and the off-board charger, which is distinct from the actual vehicle—each of which can be classified into three power levels reflecting charging characteristics shown in

Table 1. Converter specifications and requirements.

Charger Location		Level 1	Level 2	Level 3
On-board chargers	Vac	120 V	240 V	-
	Iac	12 A	80 A	-
	Plevel	1.4 kW	<19.2 kW	>20 kW
Off-board chargers	Vdc (V)	Vdc ≤ 450	Vdc ≤ 450	Vdc ≤ 600
	Idc (A)	Idc ≤ 80	Idc ≤ 200	Idc ≤ 400
	Plevel (kW)	Plevel ≤ 36	Plevel ≤ 90	Plevel ≤ 240

by the Society of Automotive Engineers (SAE) [20,21,22,23]. It can be found that the on-board charger can draw AC current from any available power outlet and can efficiently charge the batteries. However, drawbacks, including a long charging time, low efficiency, and low reliability, are apparent. In addition, the on-board charger located on the vehicles can add volume and weight. Since a normal 3.3 kW charging power requires a 6–8 h charging time, most users have no choice but to recharge overnight. The off-board charger, called a fast-charger, can directly draw power from the DC bus, which can provide sufficient power for vehicles within a short period of time. For a normal 50 kW charging power, the Nissan Leaf, with its 24 kWh battery pack, takes only half an hour to recharge. The off-board charger has certain advantages compared with the on-board charger, such as a high power density and a high efficiency. The high-frequency-isolated DC-DC converters with a high power capacity are preferable to EVs’ off-board chargers in terms of satisfying the safety requirements of charging multiple EVs within an acceptable period of time [24,25,26,27,28,29,30].

In recent years, phase-shift full-bridge (PSFB) has become a popular topology for charging, but there are some obvious defects for a traditional zero-voltage-switching (ZVS) PSFB DC-DC converter. One is that ZVS cannot be achieved under light load conditions due to the limited ZVS range for lagging-leg switches. Another defect is the excessive conduction loss caused by the primary reflected current from the output inductor current [30]. The dual active bridge (DAB) converter is also a useful topology for high-power applications. It is difficult to achieve the ZVS of all active switches in the DAB converter, as the switch stress is raised in this condition. Although some switching control strategies can improve this problem, this complex modulation increases the burden for the controller and for researchers [31,32,33].

This paper proposes a high-frequency isolation (HFI) charging DC port topology. It can serve as the interface between unipolar/bipolar DC buses and electric vehicles (EVs) through the two-power-stage system structure, which combines a front-end three-level buck converter with a back-end LLC resonant converter. The DC output voltage can be regulated within the desired voltage range by the front-end three-level buck converter. Zero-voltage switching (ZVS) for the inverter side and zero-current switching (ZCS) for the rectifier side can be realized by the back-end LLC resonant converter. In Section 2, principles of the proposed charging port topology are explored in detail. In Section 3, the features and characteristics are analyzed. In Section 4, the design conditions of the three-level output filter and high-frequency isolation transformer are explored. In Section 5, a 20 kW prototype is designed and tested. The experimental results are presented to verify the validity and performance of the proposed fast-charging DC port system structure.

2. High-Frequency-Isolation Charging Port Topology

The structure of the proposed two-power-stage system for the EV’s HFI charging port is shown in Figure 2. It combines the front-end three-level buck converter, which has the ability of regulating the output DC voltage, with the back-end LLC resonant converter, which experiences high-frequency electrical isolation. Additionally, the input terminals (T, M, B) of the three-level converter can flexibly fit the bipolar DC bus or unipolar DC bus.

The main features of the system are illustrated as follows:

(1)

According to the three-level converter, the low-voltage level rating power switches can be adapted/selected for the half-voltage stress of total DC bus. Additionally, the proposed system structure can be practical under high-voltage and high-power conditions.

(2)

The PWM phase-shift control for the front-end three-level buck converter can double the equivalent switching to greatly reduce the intermediate output LC filter volume.

(3)

The proposed structure can regulate the DC power balance without extra balancing circuits when the input terminals (T, M, B) are interfaced with the bipolar DC bus.

(4)

The back-end LLC resonant converters have the advance characteristics of zero-voltage switching (ZVS) of the inverter side and zero-current switching (ZCS) of the rectifier side. In particular, it can achieve better performance under quasi-resonant frequency mode, which greatly decreases the loss of switches.

(5)

The magnetizing current of LLC high-frequency transformer can be automatically modified by following the different DC output voltages, which have the self-adaption ZVS condition for decreasing the circulating current.

2.1. Front-End Three-Level Buck Converter

Figure 3 shows the circuit diagram of the front-end three-level buck converter, which consists of two power switches (Q₁, Q₂) with the freewheeling diodes (D_s1, D_s2), the split DC-link capacitors (C_in1, C_in2), and the intermediate LC filter (L_f1, L_f2, C_f).

2.1.1. Operating Principle

As shown in Figure 3, two switches (Q₁, Q₂) of the three-level buck converter are modulated by the PWM phase-shift control with the same duty cycle D. Figure 4a shows the converter operating waveforms for D < 0.5. Figure 4b shows the operating waveforms for D = 0.5, and Figure 4c shows the operating waveforms for D > 0.5. It can be found in Figure 4 that the equivalent frequency of filter inductor current i_L is twice the device switching frequency. When D = 0.5, this feature can reduce the output filter volume when the current ripple Δi_L for the filter inductor is nearly zero.

Figure 5 shows the four operating stages’ ((A) t₀–t₁, (B) t₁–t₂, (C) t₂–t₃, (D) t₃–t₄) equivalent circuits for D > 0.5. It can be found that there are three switching modes:

Mode 1:

When Q₁ is turned on and Q₂ is turned off, the generated voltage V_AB is half of the DC side voltage 0.5 V_dc;

Mode 2:

When Q₁ and Q₂ are turned on at the same time, the generated voltage V_AB is the total DC side voltage V_dc;

Mode 3:

When Q₁ is tuned off and Q₂ is turned on, the generated voltage V_AB is half the DC side voltage 0.5 V_dc.

Figure 6 shows the equivalent circuits of two stages ((A) t₀–t₁, (B) t₁–t₂) in the half cycle for D = 0.5. Q₁ and Q₂ are conducting alternatively. Hence, the generated voltage V_AB is half of the DC side voltage 0.5 V_dc.

Figure 7 shows the four operating stages ((A) t₀–t₁, (B) t₁–t₂, (C) t₂–t₃, (D) t₃–t₄) equivalent circuits for D < 0.5. The three models are as follows:

Mode 1:

When Q₁ is turned off and Q₂ is turned off at the same time, the generated voltage V_AB is 0;

Mode 2:

When Q₁ is turned off and Q₂ is turned on, the generated voltage V_AB is half of the DC side voltage 0.5 V_dc;

Mode 3:

When switch Q₁ is turned on and Q₂ is turned off, the generated voltage V_AB is half of the DC side voltage 0.5 V_dc.

2.1.2. The Inductor Current Ripple Analysis

The converter output inductor current ripple Δi_L is derived for D > 0.5 and D < 0.5, respectively, through the operating waveforms, as shown in Figure 4 [34,35].

$$\Delta i_{\mathrm{L}}=\{\begin{array}{l}\frac{(1-D)(2D-1)V_{\mathrm{dc}}T}{2(L_1-L_2)}\hspace{1em}D\ge 0.5\\ \frac{(1-2D)V_{\mathrm{dc}}}{2(L_1+L_2)}DT\hspace{1em}D<0.5\end{array}$$

(1)

where T is the switching period, and D is the duty cycle.

Thus, the maximum output inductor current ripple Δi_Lmax can be expressed as follows:

$$\Delta i_{\mathrm{Lmax}}=\frac{V_{\mathrm{dc}}T}{16(L_1+L_2)}$$

(2)

Figure 8 shows the graph for the relation between the duty cycle D and the output inductor current ripple Δi_L. As can be seen, the output inductor current ripple can reach the peak value Δi_Lmax for D = 0.75 and D = 0.25, while the output inductor current ripple Δi_L is nearly zero for D = 0.5.

2.2. Back-End HFI LLC Resonant Converter

The structure of the back-end LLC resonant converter is demonstrated in Figure 9. The structure consists of the full-bridge inverter, a resonant tank, and a rectifier, which can achieve the high-frequency isolation (HFI) between the common DC bus and the EVs. The full-bridge inverter contains two parallel legs with four power switches S₁–S₄ including their anti-parallel diodes D₁–D₄ and parallel capacitors C₁–C₄. Meanwhile, the resonant tank is composed of the resonant capacitor C_r, the series resonant inductor L_r, and the high-frequency isolation (HFI) transformer, while the rectifier is composed of four diodes D_R1–D_R4. The resonant inductor L_r can be replaced by the leakage inductor of the transformer. The resonant capacitor C_r can also filter the DC component and prevent DC magnetic bias. The LLC resonant converters have advance characteristics of zero-voltage switching (ZVS) of inverter side and zero-current switching (ZCS) of the rectifier side. In particular, it can achieve better performance under quasi-resonant frequency mode and greatly reduce switching losses and improving system efficiency. The features of the back-end LLC resonant converter stated above have great advantages for power electronics transformer (PET) applications under high-voltage and high-power conditions.

2.2.1. Operating Principle

The operating principle and the key waveforms of the proposed back-end LLC resonant is shown in Figure 10, including the gate signals of active switches (S₁–S₄), the series resonant inductor current i_pri, the magnetizing inductor current i_m, and the output current i_sec on the secondary side.

As for the resonant circuit, there are eight operation modes dependent on the direction of the primary current i_pri and the inverter working conditions.

(1) Mode 1 (t₀–t₁)

This mode begins when S₂ and S₃ are turned off at t₀, when the resonant inductor L_r current i_pri flows in the negative direction. Meanwhile, the C₁ and C₄ discharge until V_ce1 and V_ce4 reach zero and the primary current i_pri decreases. In this period, the magnetizing inductor current i_m is equivalent to i_pri; thus, there is no induced current on the secondary side. The stored energy acquired by the output capacity C_O is transferred to the load.

(2) Mode 2 (t₁–t₂)

In this mode, the i_pri is still negative and will flow via the anti-parallel diodes of D₁ and D₄. The ZVS condition is achieved for S₁ and S₄. The magnetizing inductor current i_m increases linearly when i_sec begins to increase.

(3) Mode 3 (t₂–t₃)

This mode begins when the resonant inductor current i_pri becomes positive. Now, the magnetizing inductor current i_m linearly increases with the resonant L_r while the resonant capacitor C_r works in a series-resonant condition. This mode ends when i_m and i_pri are equal at t₃. At the same time, the output current i_sec decays to zero. Both rectifier diodes D_R1 and D_R4 are turned off and ZCS is achieved.

(4) Mode 4 (t₃–t₄)

This mode begins when the resonant inductor current i_pri is equal to the magnetizing inductor current i_m at t₃, and both of them increase linearly while the i_sec is equal to zero. In this mode, the output is separated from the high-frequency transformer and the stored energy acquired by the output capacity C_O is transferred to the load. This mode ends when S₁ and S₄ are turned off at t₄.

For the next half cycle, the operation is opposite to the analysis above.

2.2.2. Voltage Gain Characteristics of the LLC Resonant Converter

As can be seen from Figure 11, the equivalent circuit diagram works under quasi-resonant frequency, in which R_eq is equivalent to the AC load of DC load R. The output DC load R is replaced by the equivalent AC load R_eq expressed as follows [36]:

$$R_{\mathrm{eq}}=\frac{8n^{2}R}{{\pi }^2}$$

(3)

where n is the transformer turn ratio.

To simplify the analysis, the input voltage can be equivalent to a square-wave voltage source U_square, which needs to be Fourier-transformed:

$$U_{\mathrm{square}}(\omega t)={\displaystyle \sum _{k=1}^{\infty }{u}_{\sin }(k\omega t)}=\frac{4U_{\mathrm{m}}}{\pi }(\sin \omega t+\frac{1}{3}\sin 3\omega t+\cdots )$$

(4)

Sinusoidal steady-state analysis is required for the equivalent electrical circuit. The output voltage ${\dot{U}}_{\mathrm{kout}}$ under the $k\omega$ frequency condition can be obtained by

$${\dot{U}}_{\mathrm{kout}}=\frac{{\dot{U}}_{\mathrm{m}}(x_{\mathrm{Lm}}//{\mathrm{R}}_{\mathrm{eq}})}{X_{\mathrm{Cr}}+X_{\mathrm{Lr}}+(X_{\mathrm{Lm}}//{\mathrm{R}}_{\mathrm{eq}})}=\frac{U_{\sin }k\omega \frac{jk\omega L_{\mathrm{m}}{\mathrm{R}}_{\mathrm{eq}}}{jk\omega L_{\mathrm{m}}+{\mathrm{R}}_{\mathrm{eq}}}}{\frac{1}{jk\omega C_{\mathrm{r}}}+jk\omega L_{\mathrm{r}}+\frac{jk\omega L_{\mathrm{m}}{\mathrm{R}}_{\mathrm{eq}}}{jk\omega L_{\mathrm{m}}+{\mathrm{R}}_{\mathrm{eq}}}}$$

(5)

Then, the instantaneous value of the output voltage $u_{\mathrm{out}}(t)$ is given by

$$u_{\mathrm{out}}(t)={\displaystyle \sum _{k=1}^{\infty }{u}_{\mathrm{kout}}(k\omega t)}$$

(6)

The RMS of output voltage U_out yields

$$U_{\mathrm{out}}=\sqrt{U_{1\mathrm{out}}^2+U_{2\mathrm{out}}^2+U_{3\mathrm{out}}^2+\cdots }$$

(7)

Hence, the LLC resonant converter voltage gain G is

$$G=\frac{U_{\mathrm{out}}}{U_{\mathrm{m}}}$$

(8)

where U_m is the RMS of the square-wave voltage source.

The value of L_r is the intrinsic parameter of the transformer, and the values of L_m and C_r are calculated by Equation (11). An LLC resonant converter with a 44 kHz resonant frequency has been designed, and the key parameters are L_r = 9.7 μH, L_m = 230 μH, and C_r = 1.32 μH. The experimental voltage gain characteristics of the LLC resonant converter under different load scenarios are shown in Figure 12 through the frequency response analyzer (VENABLE Model 3120). As can be observed in Figure 12, the LLC resonant converter shows the voltage gain characteristics, which are almost independent of the load when the switching frequency f_s is around the resonant frequency f_r called a quasi-resonant frequency. This is a distinct advantage of LLC resonant converter compared to the other DC-DC converters.

3. Features and Characteristics

3.1. Three-Level Buck Converter Working at Higher DC Modulation Index

The three-level buck converter is placed in front of the LLC resonant converter, which can help to obtain the desired output voltage with the higher modulation index D₁ expressed in Equation (9) comparison to the other DC-DC converters:

$$D_1=\frac{V_{\mathrm{o}}}{V_{\mathrm{dc}}}n$$

(9)

The DC modulation index D₂ of the traditional DC converter given by

$$D_2=\frac{V_{\mathrm{o}}}{V_{\mathrm{dc}}}$$

(10)

where V_o is the output voltage, V_dc is the input voltage, and n is the transformer turn ratio. A step-down transformer (n > 1) has been used; hence, D₁ > D₂.

Taking the input voltage V_dc = 760 V and the output voltage V_o range from 200 to 500 V of the HFI DC–DC converter as an example. If n = 760/500, after the aforementioned calculation, the range of DC modulation index D₁ is from 0.4 to 1, while the range of the DC modulation index D₂ is from 0.26 to 0.65. Thus, the front-end three-level buck converter working at higher DC modulation index increases the voltage utilization ratio and improves efficiency.

3.2. The Proposed DC Charging Port Having Different Power Balancing Capability for a Bipolar DC Bus

The split DC-link capacitors (C_in1, C_in2) connected on three-level buck converter are used as the interface between the bipolar DC buses. If there is a power imbalance in the bipolar DC bus, changing the DC modulation index of the upper and lower legs can balance the power between the DC buses. Figure 13 shows a diagram of power balance management. If the power on the positive DC side is higher than the negative DC side, it is necessary to increase the DC modulation index of the upper leg and reduce the DC modulation index of the lower leg. Therefore, the fast charger discharges more power from the positive to balance the bipolar DC bus. Alternatively, if the power on the positive DC side is less than the power on the negative DC side, it can balance the bipolar DC bus by adjusting the modulation index of the upper and lower legs.

3.3. LLC Resonant Converter Having ZVS Conditions with Different Magnetizing Current at Different Output Voltages

There is a condition for achieving ZVS:

The energy stored in the magnetizing inductor is larger than the energy stored in the switch parallel capacitance. According to [37], a generalized expression for magnetizing inductance is derived as

$$L_{\mathrm{m}}<\frac{v_{\mathrm{in}}}{N{v}_{\mathrm{out}}}\cdot \frac{t_{\mathrm{d}}}{16C_{1}f}$$

(11)

where t_d is the dead-time interval, N is the ratio of transformer, and $C_1$ is the parasitic capacitance.

The structure of the proposed HFI charging port is shown in Figure 2. When the DC modulation index of front-end three-level buck converter was charged, the middle voltage V_m will also be changed, causing the magnetizing inductor current i_m to change. The simulation waveforms of magnetizing inductor i_m under different inverter output voltages (V_CD) are shown in Figure 14. It can be clearly seen that i_m is changing with V_CD, but the ZVS can be achieved by the magnetizing inductor L_m by drawing energy from the parallel capacitors C₁–C₄ until the energy stored in the parallel capacitance discharged thoroughly. This is because the voltage of parallel capacitor V_cex decreases to zero during the dead time, as the converter has the self-adaption ZVS condition.

3.4. LLC Resonant Converter Containing Constant Current at Different Output Voltages with the Same Charging Current Command

As the proposed back-end LLC resonant converter directly transfers power to the EV’s batteries, the output current remains constant whenever the output voltage changes in constant-current charging mode. The battery terminal voltage increases when the output current remains constant, while the input voltage V_m of LLC resonant converter increases along with the output voltage V_o, causing the magnetizing inductor current i_m to also increase. In order to maintain the output current i_o constant following the output voltage increasing, the resonant current i_pri increases a little along with the magnetizing inductor current i_m. Therefore, the resonant current waveform is not affected by the output voltage V_o. On the contrary, if the LLC resonant converter is placed at the front end and is connected directly to the DC bus, the LLC input voltage V_in is clamped at the DC bus voltage V_dc. Thus, it keeps constant and the waveform of magnetizing inductor current is also constant. In this condition, the input current inevitably changes with the output voltage fluctuates in constant-current charging mode. Thus, the whole resonant process could not be optimized in different power conditions.

4. Design Conditions

4.1. Front-End Three-Level Buck Converter Inductor Filter Design

The maximum value of the filter inductor current ripple Δi_Lmax can typically be 10–30% of the peak current:

$$\Delta i_{\mathrm{L}\max }=\frac{V_{\mathrm{dc}}T}{16\left(L_1+L_2\right)}$$

(12)

Thus, the total filter inductor L_f can be derived as

$$L_{\mathrm{f}}=L_1+L_2=\frac{V_{\mathrm{dc}}T}{16\Delta I_{\mathrm{L}\max }}$$

(13)

The Δi_L reaches the maximum value for the duty cycle D = 0.25 and D = 0.75. A 20 kW prototype with a 760 V input DC voltage and a 200–500 V output DC voltage with a working frequency of f_s = 20 kHz was built and tested. Based on the calculation above ΔI_Lmax = 6 A, the filter inductor L₁ = L₂ = 200 μH can be derived. The next step is to select a suitable magnetic core and calculate the desirable turns N.

The front-end three-level buck converter can increase the equivalent switching twice using the PWM phase-shift control, so the frequency of inductor current ripple is 40 kHz. The formula of the Kool Mu window area W_a is derived as follows [38]:

$$W_{\mathrm{a}}{A}_{\mathrm{e}}\ge \frac{L{I}_{\mathrm{rms}}{I}_{\mathrm{pk}}}{J{B}_{\mathrm{m}}{K}_{\mathrm{u}}}{10}^4$$

(14)

where the utilization rate of the window K_u is 0.5, and the maximum magnetic flux density B_m is 0.45 T. The current density J is 400 A/cm². A_ε is the cross area of the wire.

Hence, according to W_a1A_e1 > 341 cm⁴, W_a1A_e1 = 350 cm⁴ is better for designing a necessary inductor in the Power Electronics Laboratory based on actual engineering problems. A magnetic core K8020E060 was selected, of which the window area W_a2 is 11.1 cm², and the cross-sectional area A_e2 is 3.89 cm². The number of magnetic core S can be shown to be

$$S=W_{\mathrm{a}1}{A}_{\mathrm{e}1}/W_{\mathrm{a}2}{A}_{\mathrm{e}2}$$

(15)

Based on Equation (15), the number S is 8. The inductor L can be denoted as

$$L=S{A}_{\mathrm{L}}{N}^2$$

(16)

where A_L is 190 nH; thus, the number of turns N = 17.

The experimental filter inductor prototype in accordance with the theoretical analysis mentioned above is presented in Figure 15.

4.2. Back-End LLC Converter ZVS Condition

For the back-end LLC resonant converter switches (S₁–S₄), the ZVS condition can satisfy from zero to full-load conditions because the magnetizing current i_m shown in Equation (17) is independent of the loads:

$$i_{\mathrm{m}}=\frac{V_{\mathrm{in}}}{4L_{\mathrm{m}}{f}_{\mathrm{s}}}$$

(17)

The input voltage V_in of the LLC resonant converter is clamped to the front-end buck converter regulated by the DC modulation index. The energy stored in the magnetizing inductor is sufficient for achieving ZVS for switches (S₁–S₄) within the voltage range. In this condition, another key point is regulating the dead time to satisfy the condition under which the energy stored in parallel capacitance can decay thoroughly.

4.3. High Frequency LLC Transformer Design

The high-frequency electrical isolation can be realized by the high-frequency transformer, which is the main component of the back-end LLC resonant converter with the bus converter function. The turn ratio n and the number of turns N are the key parameters for designing a high-frequency transformer. A 20 kW prototype with a 760 V input and a 200–500 V output DC voltages with a working frequency f_s = 40 kHz was designed, so the turn ratio n (760/500) was adopted. The peak value of input voltage E₁ can be calculated as follows:

$$E_1=4f{N}_1{B}_{\mathrm{m}}{A}_{\mathrm{core}}$$

(18)

where N₁ is number of turns on the primary side, A_core is the cross-sectional area of the main magnetic circuit of the magnetic core, B_m is the maximum flux density, and f is the working frequency.

Thus, the number of turns on the secondary side N₂ can be obtained:

$$n=\frac{N_1}{N_2}$$

(19)

The experimental high-frequency LLC transformer is presented in Figure 16.

5. Experimental Results

The 20 kW prototype with an input voltage of 760 V and an 200–500 V output voltage, which integrates with the front-end three-level buck converter working at a frequency of 20 kHz and the back-end LLC resonant converter working at a frequency 40 kHz, was designed and tested. The experimental results are presented here to verify the validity and performance of the proposed fast-charging DC port system structure.

The photograph of the hardware prototype is shown in Figure 17, and the specifications of the prototype are shown in

Table 2. Circuit parameters of the HFI fast-charging DC port system.

Description	Parameters
Input voltage Vdc	760 V (DC)
Output voltage Vo	200–500 V (DC)
Transformer turn ratio n	17:12
Three-level buck converter working frequency f	20 kHz
LLC resonant converter working frequency fs	40 kHz
Resonant frequency fr	44 kHz

.

In the power electronics laboratory, the ITEM number of the resonant capacitor is C4BSYBX3330Z_F_, the manufacturer is KEMET, U_n = 3000 V, and the value is 0.33 $\mathsf{\mu }\mathrm{F}$. The ITEM number of the rectifier diode is DSEI2X101-12A, and the manufacturer is IXYS. The ITEM number of the IGBT is SKM200GB125D, and the manufacturer is SEMIKRON INTERNATIONAL. The ITEM number of the voltage capacitance is SHK-700-540-FS, and the manufacturer is EACO. The ITEM number of the transformer core is K8020E060.

Figure 18 shows the waveforms of the proposed HFI charging DC port with D = 0.75; the total input voltage V_dc = 760 V, the LLC resonant converter input voltage V_in = 570 V, and the output voltage V_o = 400 V; the total input current i_dc = 26 A, the inductor current i_L = 35 A, and the output current i_o = 50 A with the resistive DC load R₀ = 8 Ω; the output power is about 20 kW. It can be seen that the system performs well while working in the steady-state operation and the experimental results are consistent with the theoretical analysis above.

5.1. In Steady-State Operation

Figure 19 shows the experimental waveforms of the front-end three-level buck converter output voltage V_AB and inductor current i_L under different DC modulation indices (D = 0.4, D = 0.5, D = 0.75). The input voltage V_dc = 760 V and the load R₀ = 8 Ω. As seen from Figure 20a, the V_AB voltage is changed between 0 (Q₁ and Q₂ are off-state, D₁ and D₂ are on-state) and 380 V (Q₁ and D₂ are on-state or Q₂ and D₁ are on-state) for D = 0.4 and the output power is about 5.7 kW. The inductor current ripple Δi_L = 3.8 A is very close to the result of Equation (1). From Figure 20b, V_AB is fixed at 380 V (Q₁ and D₂ are on-state or Q₂ and D₁ are on-state), and the inductor current ripple Δi_L almost reaches zero. This is because there is no voltage ripple on the filter inductors with D = 0.5, and the output power is nearly 9 kW. As can be seen from Figure 20c, V_AB is changed between 380 V (Q₁ and D₂ are on-state or Q₂ and D₁ are on-state) and 760 V (both Q₁ and Q₂ are on-state) for D = 0.75 with about 20 kW of the output power. It can be found that the value of the output inductor current ripple reaches the maximum value Δi_Lmax, which is about 6 A.

Figure 20 shows the experimental waveforms of the intermediate filter inductor current i_L, the LLC primary side current i_pri, the LLC secondary side current i_sec, and the LLC inverter-side output voltages (V_CD) under different front-end modulation indices (with D = 0.4 having an output power of about 5.7 kW, D = 0.5 having an output power of about 9 kW, and D = 0.75 having an output power of about 20 kW). The input voltage, V_dc = 760 V, and the load, R₀ = 8 Ω. It can be found that the primary side current i_pri, secondary side current i_sec, and inverter-side output voltages V_CD are changed due to different modulation indices. Additionally, the LLC resonant converter has good characteristics of zero-voltage switching (ZVS) with a different magnetizing inductor current i_m. The current waveforms on the primary side and secondary side present smooth curves that are consistent with the simulation results; thus, the viability and performance of the two power-stage system structure has been proved.

The ZVZCS characteristic curves of the back-end HFI LLC resonant converter under different modulation indices are shown in detail in Figure 21. The input voltage V_CD = 760 V and the load, R₀ = 8 Ω. V_ce1 is the voltage of switch S₁ and V_gs1 is the gate signal; i_pri is the primary side current and i_sec is the secondary side current. It can be seen from Figure 21 that V_gs1 begins to turn on switch S₁ after the V_ce1 decreases to zero; thus, ZVS can be achieved from zero to full-load conditions. Additionally, ZCS can be achieved when i_sec reaches zero at the end of the resonant process under different output voltages conditions.

Figure 22 shows the experimental results of magnetizing inductor i_m current under different modulation indices D in detail under different power conditions. When the resonant process of C_r and L_r ends, the LLC primary-side i_pri is the same as the magnetizing inductor current i_m. It can be seen that i_m is changing with the inverter-side output voltage V_CD, the ZVS can be achieved adaptively and the circulating current is also decreased.

5.2. In the Load-Step Operation

In order to verify the viability and performance of the fast-charging DC port, dynamic-state experiments have been designed with changing loads. Figure 23 and Figure 24 show the transient experimental waveforms of the front-end three-level buck converter output voltage V_AB, the inductor current i_L, the LLC inverter-side output voltage V_CD, and the primary side current i_pri during a load step-up of the resistive load from 16 to 27 Ω and a load step-down of the resistive load from 27 to 16 Ω. As can be seen, the output voltages V_AB and inverter output voltage V_CD are independent of the resistive load, while the inductor current i_L and the primary side current i_pri reach a new stable state in a short time. Experimental results have shown that the proposed system structure can perform well under the conditions of a transient operation.

Finally, the experimental results above have shown the characteristic of the front-end three-level buck converter and the back-end LLC resonant converter in a steady-state operation and in a load-step operation. The validity and performance of the proposed HFI fast-charging DC port system structure has thus been proved.

6. Conclusions

A high-frequency-isolation (HFI) charging DC port can serve as the interface between unipolar/bipolar DC buses and electric-vehicles (EVs) based on a double power-stage system structure, which combines a front-end three-level converter with a back-end LLC resonant converter. Principles of the proposed charging port topology are studied in detail, while the features and characteristics are analyzed. Then, the design conditions of the three-level output filter and high-frequency isolation transformer are explored. Finally, a 20 kW prototype is designed and tested. The experimental results are presented to verify the validity and performance of the proposed HFI fast-charging DC port system structure.