An SIW Quasi-Pyramid Horn Antenna Based on Patch Coupling Feed for Automotive Radar Sensors

Abstract

An SIW quasi-pyramidal horn antenna based on patch coupling feed with reduced machining difficulty and facilitated integration with the radar chip is proposed in this paper. Compared with the metal pyramid horn antenna, the Rogers 5880 dielectric substrate based on the SIW structure is used to form the horn structure and waveguide structure, which effectively reduces the difficulty of machining the antenna. The patch coupling feed structure provides a solution for integrating the SIW quasi-pyramid horn antenna with the radar chip. The proposed SIW quasi-pyramid horn antenna element achieves approximately 9 dBi realized gain, about 95% radiation efficiency and 8.2 GHz bandwidth (74.1–82.3 GHz). A four-port inverting power divider was designed to verify the feasibility of forming an array with antenna elements. The designed antenna array achieves approximately 14.5 dBi realized gain, about 80% radiation efficiency and 7.2 GHz bandwidth (74.3–81.5 GHz). Simulation and measurement results maintain good agreement for the antenna array. To further assess the impact of errors on the performance of the proposed antenna array, we have implemented a corresponding error analysis. The proposed antenna element and antenna array show promising potential for application in automotive radar systems.

1. Introduction

With the growing demands for the integration, modularization, and universalization of advanced driver assistance systems (ADAS), automotive radar sensors are evolving towards miniaturization, integration, and affordability [1,2,3,4,5]. Typically, antennas play a crucial role in automotive radar sensors. Therefore, the evolution of automotive radar sensors necessitates antennas with the characteristics of miniaturization, low-cost, and easy integration. Furthermore, antennas employing automotive radar sensors can also be applied in fields such as human posture recognition [6] and vital signs detection [7].

In [8,9,10,11,12], microstrip patch antennas and microstrip comb antennas are extensively utilized in automotive radars due to their easy integration capabilities with 77 GHz radar chips. In [2], the picture of the microstrip antenna directly integrated with a radar chip is presented. However, the gain and radiation efficiency of a single microstrip patch and comb antenna element are relatively low, posing significant challenges in meeting the requirements for spatial resolution, imaging quality, and imaging accuracy of automotive radar [13,14,15].

Achieving high-precision imaging and improving point cloud quality require high gain and high radiation efficiency [16,17]. As illustrated in Figure 1, the antenna serves as one of the pivotal components of the automotive radar sensor. When the information parameters of the detected target are consistent, the use of high gain and high efficiency antennas can alleviate the burden of back-end data processing, thereby enhancing the quality and precision of imaging. Therefore, it is very crucial to design an antenna with high efficiency and high gain that can be applied to automotive radar sensors, which is of great significance to improving the driving safety of the car and safeguarding the lives of drivers. The metal pyramid horn antenna exhibits high gain and radiation efficiency, making it commonly utilized as the feed source for reflector antennas or as an array element in array antennas. Furthermore, it is worth considering the application of metal pyramid horn antennas in automotive radars. However, the application of metal pyramid horn antennas presents several challenges such as the complex machining of high frequency metal structures, increased machining costs, and the difficulty of integration with radar chips [18].

In this paper, an SIW quasi-pyramid horn antenna based on patch coupling feed is proposed to reduce the machining cost of the antenna and facilitate integration between the horn antenna and the radar chip. The proposed antenna is characterized by high gain and high radiation efficiency, and can effectively improve imaging quality and imaging accuracy when applied to automotive radar sensors. Among them, the side walls of the horn structure and waveguide structure are composed of SIW structures, which helps to alleviate the complexity of antenna fabrication. The patch coupling feed structure serves as a “bridge” connecting the SIW quasi-pyramid horn antenna and the radar chip, offering a solution for integrating the horn antenna and the radar chip. Furthermore, the feasibility of forming an array using the proposed SIW quasi-pyramid horn antenna element and integrating it with the radar chip has been verified.

2. Antenna Element Design and Analysis

2.1. SIW Quasi-Pyramid Horn Antenna Element

An exploded view of the proposed SIW quasi-pyramid horn antenna is shown in Figure 2a. The overall structure of the antenna element consists of eight layers. Each layer employs a Rogers 5880 dielectric substrate (${\epsilon }_r$ = 2.2, h = 0.254 mm). To minimize the interference (including propagation loss) of the substrate on electromagnetic waves, a rectangular through-hole structure is positioned in the center of the rectangular SIW structure; the distance d_i from the edge of the rectangular through-hole structure to the edge of the rectangular SIW structure is set to 0.01 mm. Moreover, the d_i value of each layer of dielectric substrate remains consistent. A side view of the antenna element is shown in Figure 2b. It can be divided into two main structures: the horn and the waveguide. The side walls of the horn structure are stepped. In Figure 2c, the SIW rectangular structure of each layer is presented in detail. According to the design guideline of the SIW structure [19,20], the diameter of the metallized via hole has been determined to be 0.4 mm, with the distance between adjacent holes set at approximately 0.6 mm. As depicted in Figure 2b, the aperture size of the pyramid horn is determined based on the optimal gain design principles outlined in [21]. After determining the aperture size of the horn, the length b_i in the E-plane and width a_i in the H-plane of the rectangular SIW structure for each layer are determined accordingly. The long and wide sides of the rectangular SIW structure of each layer of the horn structure and the spacing between adjacent metallized vias are represented by L_i, W_i and s_i, respectively. Among them, b_i corresponds to the long side L_i of the rectangular SIW structure of each layer; a_i corresponds to the wide side W_i of the rectangular structure of each layer. The long and wide sides of the rectangular SIW structure of the waveguide structure and the spacing between adjacent metal vias are represented by L, W, and s, respectively. The design process for determining the number of metallized vias corresponding to the length b_i and the spacing s_i between adjacent metallized vias is as follows (the same design process applies to width a_i):

• Calculate the number Nas_i of approximate spacing between adjacent metallized vias. b_i ÷ 0.6$=$Nas_i. The purpose of dividing 0.6 is to ensure that the distance between adjacent metallized via holes is approximately 0.6 mm, thereby satisfying electromagnetic wave constraints. Nas_i is generally rounded off to the nearest whole number.
• Determine the number N of metallized vias. N$=$Nas_i + 1.
• Determine the spacing s_i between adjacent metallized vias. s_i$=$b_i ÷ N. For varying length b_i, s_i is a variable value. However, the value of s_i is always maintained at approximately 0.6mm.
• Finally, based on the aforementioned design steps, the number N of metallized vias and the spacing s_i between adjacent metallized vias can be obtained.

Further, the detailed design parameters of the proposed SIW quasi-pyramid horn antenna are presented in

Table 1. Parameters of the Proposed SIW Quasi-Pyramid Horn Antenna.

Parameters	L8 (b8)	L7 (b7)	L6 (b6)	L5 (b5)	L4 (b4)	L
Value (mm)	4.94	4.54	4.14	3.74	3.34	3.34
Parameters	W8 (a8)	W7 (a7)	W6 (a6)	W5 (a5)	W4 (a4)	W
Value (mm)	3.67	3.27	2.87	2.47	2.07	2.07
Parameters	s8	s7	s6	s5	s4	s
Value (mm)	0.618	0.649	0.591	0.623	0.668	0.668

.

2.2. Simulation Performance Analysis

The structure involved in this paper is simulated and optimized in HFSS software. The reflection coefficient and realized gain of the proposed SIW quasi-pyramid horn antenna element are presented in Figure 3. The radiation pattern of the proposed SIW quasi-pyramid horn antenna element based on WR10 standard waveguide feeding at the 77 GHz frequency point is shown in Figure 4. The E-plane 3 dB beamwidth of the antenna element radiation pattern is approximately 40°. The H-plane 3 dB beamwidth of the antenna element radiation pattern is approximately 50°. Furthermore, according to the 3 dB beamwidth of the radiation pattern, the proposed SIW quasi-pyramid horn antenna element can be applied to vehicle-mounted millimeter wave short-range radar. In order to further assess the performance of the proposed antenna, we will compare it with a metal pyramid horn antenna of identical dimensions. The comparison results are depicted in Figure 3.

The electric field distributions of the proposed antenna at 76 GHz, 77 GHz, 79 GHz, and 81 GHz frequency points are illustrated in Figure 5. It can be observed from Figure 5 that at the crucial frequency points of the automotive radar, the electromagnetic waves are confined around the metallized vias, and no leakage occurs. This suggests that the proposed antenna structure and dimensions are highly compatible with various operating frequency points.

The impact of the number of horn layers on the reflection coefficient and realized gain is illustrated in Figure 6. Based on Figure 6a, the optimal impedance matching between the antenna element and free space is observed when the horn structure consists of five layers. In Figure 6b, as the number of horn layers increases, the average gain (72–82 GHz) gradually rises. The number of layers in the horn structure can be adjusted according to the specific gain requirements.

The resonant frequency of the proposed antenna can be affected by the aperture size of the waveguide part in Figure 2c. The effect of aperture size on the resonant frequency of the proposed antenna element is shown in Figure 7.

3. Array Design

3.1. Patch Coupling Feed Structure Design

To address the integration problem between the quasi-pyramid horn antenna and the radar chip, a patch coupling feed structure is proposed. The overall structure of the quasi-pyramid horn antenna and patch coupling feed structure is shown in Figure 8. Antenna 1 and Antenna 2 are shown separately in Figure 8a. Antenna 1 does not add a conversion structure and utilizes wave port excitation. Antenna 2 incorporates an SIW to microstrip structure and utilizes lumped port excitation. Among them, the antenna element fed by the wave port can be connected to the SIW power divider, facilitating the formation of an array design. Meanwhile, the antenna element fed by the lumped port can be directly connected to the radar chip, enabling integrated design. The integrated design scheme of the antenna element and radar chip is verified in Figure 8b after the SIW structure is converted to a microstrip line. The patch coupling feed structure model is shown in Figure 9. As depicted in Figure 8a and Figure 9, the patch coupling feed structure consists of a Rogers 5800 dielectric substrate with a metal layer covering both the upper and lower surfaces of the dielectric substrate. The metal layer consists of a copper layer (0.035 mm) and a gold layer (0.05 mm) that is plated onto the copper layer. The detailed design parameters of the patch coupling feed structure are presented in

Table 2. Parameters of the Designed Patch Coupling Feed Structure.

Parameters	CL	CW	SL	SW	PL	PW	a	CPL	CPW
Value (mm)	3.99	2.75	2.1	1.25	1.86	0.94	2.4	1.025	0.945

. The reflection coefficient and realized gain of the SIW quasi-pyramid horn antenna 1 and antenna 2 based on the patch coupling feed structure are shown in Figure 10b. The electric field distribution diagram of antenna 1 is shown in Figure 10c. It can be observed from Figure 10a that compared to the WR10 waveguide feed, the patch coupling feed structure exhibits a narrower bandwidth (still covers 76–81 GHz). This is attributed to the relatively wide bandwidth of the WR10 standard waveguide. Simultaneously, the WR10 standard waveguide is commonly employed in measuring microwave devices operating within the frequency range from 73.8 GHz to 112 GHz. This indicates that it exhibits exceptional transmission capabilities within the 73.8 GHz to 112 GHz frequency range. Consequently, the bandwidth performance of the pyramid horn antenna is enhanced when it is fed by the WR10 standard waveguide. However, the patch coupling feed structure is primarily composed of an SIW structure. To ensure electromagnetic waves are efficiently contained within the SIW structure, it is typically essential to determine the diameter, spacing, and width of the metallized vias based on the operating frequency [19]. Increasing the spacing excessively between metallized vias can lead to leakage of electromagnetic waves, consequently diminishing the transmission performance of the SIW structure. Reducing the spacing between metallized through-holes excessively can complicate the manufacturing process. Consequently, deciding on the optimal spacing for metallized vias should take into account not only the electromagnetic wave transmission efficiency of SIW structure but also the structural strength and processing difficulty of the dielectric substrate. When the diameter, spacing and width of the metalized vias are determined, the operating frequency range for the SIW structure is consequently determined. Moreover, when employing a patch coupling feed structure in the pyramidal horn antenna, its bandwidth capabilities are constrained by the parameters of the SIW structure, resulting in a comparatively narrow bandwidth. To summarize, for a pyramid horn antenna of identical design, employing the WR10 standard waveguide feed results in superior bandwidth performance compared to utilizing the patch coupling feed structure. As a result, the bandwidth of the pyramid horn antenna is constrained by its patch coupling feed structure. Figure 10c shows that the electromagnetic energy from the feed structure is coupled to the quasi-pyramid horn antenna through the rectangular patch on the SIW resonant cavity. Among them, the electromagnetic energy is primarily concentrated at the edge of the rectangular patch. Meanwhile, in order to ensure the uniform coupling of electromagnetic energy from the radiation patch to the quasi-pyramid horn structure, it is necessary to align the center of the radiation patch with the center of the pyramid horn structure. The simulated radiation efficiencies of the antenna 1 at 76 GHz, 77 GHz, 78 GHz, and 80 GHz are 95.4%, 95.6%, 95.4%, and 94%, respectively. The simulated radiation efficiencies of the antenna 2 at 76 GHz, 77 GHz, 78 GHz, and 80 GHz are 95.6%, 95.3%, 95.4%, and 93.8%, respectively.

3.2. Design of Four-Port Inverting Power Divider

To verify the feasibility of forming an array of SIW quasi-pyramid horn antenna elements based on patch coupling feeding, a four-port inverting power divider is designed for feeding the antenna array. The spacing between adjacent elements in the antenna array is determined through simulation optimization of the antenna element shown in Figure 2. After the spacing between the array elements is established, the structural dimensions of the inverting power divider is consequently determined. Once the structural dimensions of the inverting power divider have been established, it is essential to specify the performance characteristics of the inverting power divider. Initially, the two-element antenna array structural model based on patch coupling feed is employed to ascertain the phase characteristics of the four-port inverting power divider. The two-element antenna array structural model based on patch coupling feed is shown in Figure 11. As illustrated in Figure 11, due to the positioning of the two feed ports (port 1 and port 2) in opposite directions, the in-phase radiation of the two-element antenna array can be achieved through anti-phase excitation of the two feed ports. The two feed ports are excited in anti-phase, meaning there exists a phase difference of 180° between the two feed ports. In the HFSS software, two approaches are available to achieve a 180° phase difference between the two feed ports. One approach is to set the direction of the port integrating line to the opposite direction when setting up the wave port excitation. Another approach is to set the phase difference between the two excitation ports to 180°. However, for the actual design of the power divider, the only feasible method is to set the phase difference between the two excitation ports to 180°. The approach of setting the direction of the wave port integration line in opposite directions is impractical. When adopting the method of setting a 180° phase difference between the two excitation ports, the simulation results of the radiation pattern for the two-element antenna array structural model based on patch coupling feed are shown in Figure 12. It can be seen from Figure 12 that the antenna array exhibits excellent radiation directivity. Based on the analysis of the two-element antenna array discussed above, for the four-element pyramidal horn antenna array, the phase difference between the left and right sides of the four-port inverting power divider should be set to 180° in order to achieve in-phase radiation of the four elements in the antenna array. In the simulation of the four-port inverting power divider, in order to reduce the complexity of the simulation process, the integral lines of the wave ports on the left and right sides of the power divider are set in opposite directions. In this case, when the phases of the four ports of the four-port inverting power divider are the same, the 180° phase difference between the left and right ports can be achieved. This approach facilitates data viewing. The side view of the four-port inverting power divider structural model is shown in Figure 13a. Furthermore, by connecting the four-port inverting power divider and the patch coupling feed structure, the feed of the antenna array can be realized. As shown in Figure 13c, the phase shift structure component is mainly used to adjust the phase difference between the left and right ports of the power divider. While adjusting the phase, the amplitude should also be finely tuned. The design requirement of the inverting power divider is to ensure that the amplitude of each output port is equal, and the phase difference between the left and right ports is 180°. The structural model of the four-port inverting power divider is shown in Figure 13. Its design parameters are shown in

Table 3. Parameters of the Proposed SIW Quasi-Pyramid Horn Antenna.

Parameters	a	RFx	RFy	IMx	IMy	AD1
Value (mm)	2.4	7.85	2.5	2.9	2.3	0.6
Parameters	AD2	PD1	PD2	PD3	PD4	PD5
Value (mm)	0.8	2.6	2.6	2.2	2.2	2.55

. The simulation results of the reflection coefficient, amplitude and phase of the four-port inverting power divider are illustrated in Figure 14, in which, within the operating frequency range, the amplitudes of the each ports are roughly equal, and the phase difference between the left and right ports is approximately 180°. This demonstrates that the designed four-port inverting power divider effectively fulfills the practical requirements of the antenna element for forming an array.

4. Experimental Results, Discussion, and Error Analysis

4.1. Experimental Results and Discussion

The simulation structural model of the designed SIW quasi-pyramid horn antenna array is depicted in Figure 15. Considering the actual machining accuracy, the value of d_i is set to 0.1 mm. It can be adjusted accordingly based on the actual machining accuracy without significantly affecting the antenna’s performance. To facilitate the measurement of the designed antenna array, the transition structure also utilizes the patch coupling feed structure. The parameters of the transition structures, different from those of the patch coupling feed in Figure 9, are CW = 4.13 mm and SW = 0.9 mm, respectively. The remaining structural parameters remain consistent. In order to simplify the manufacturing process of the antenna array and reduce the manufacturing cost of the antenna array, the various layers of dielectric substrates that constitute the pyramid horn antenna are combined together. The combined dielectric substrate is shown in Figure 16. After the processing is completed, the combined dielectric substrate should be cut along the longitudinal and transverse cutting lines to form each layer of dielectric substrate required for the SIW quasi-pyramid horn antenna array. A picture of the manufactured SIW quasi-pyramid horn antenna array is shown in Figure 17. The reflection coefficient of this antenna array was measured using an Agilent N5242 vector network analyzer. The radiation pattern is measured in a professional microwave anechoic chamber. Simulation and measurement results of reflection coefficient and realized gain are shown in Figure 18. The simulation and measurement results of the radiation pattern of the antenna array are shown in Figure 19. It can be seen from Figure 19 that the side lobes of the proposed antenna array are higher after forming the array. This can be attributed to the larger array element spacing. Typically, the pyramid horn antenna with a greater number of layers will exhibit an increased gain. However, augmenting the number of layers results in an expansion of the aperture size, which will force the array element spacing to become larger, thereby causing an increase in side lobes. While diminishing the layers of the pyramid horn antenna can indeed decrease the aperture size, this action simultaneously leads to a decline in gain. In the subsequent phase, the research will concentrate on strategies to enhance the gain of the pyramid horn antenna while simultaneously minimizing its side lobes. Due to the symmetrical configuration of the proposed SIW quasi-pyramid horn antenna array, the 3 dB beamwidths of the E-plane and H-plane of the radiation pattern are approximately equal. The 3 dB beamwidth of the E-plane and E-plane of the antenna array is approximately 20°. Furthermore, based on the 3 dB beamwidth characteristic of the antenna array radiation pattern, the proposed antenna array can be applied to vehicle-mounted millimeter-wave medium-range radar. The simulated radiation efficiencies of the proposed antenna array at 76 GHz, 77 GHz, 78 GHz, and 80 GHz are 80.9%, 80.7%, 78.9%, and 77.9%, respectively. Compared to the radiation efficiency of the antenna element, the antenna array exhibits lower radiation efficiency. This can be attributed to the limitation imposed by the inverting power divider on the radiation efficiency of the antenna array.

The comparison of the proposed SIW quasi-pyramid horn antenna array with other published antennas is presented in

Table 4. Comparison of the proposed SIW Quasi-pyramid horn antenna with other previously published antennas.

Ref.	Fre. (GHz)	Structure	Relative	Apeture	G. or R.G. *	Radiation
		Type	Bandwidth	Size (${\mathbf{\lambda }}_{\mathbf{0}}$)	(dBi)	Efficiency
[22]	18	SIW	19.89%	2.67×1.9 × 0.31	11.4	85%
[23]	40	SIW	69.7%	4.5 ′ × 3.9 × 2.5	14.8	85%
[24]	35	SIW	0.81% ′	12.4 ′ × 2.71 × 2.06	13.1 *	93.5%
[18]	57	Metal	10.53%	7.6 × 1.6 × 1.6	12 *	76%
This work	77	SIW	9.27%	6.3 × 5.8 × 0.59	14.56 *	80.7%

The values with “′” are approximate value based on the picture provided in the relevant research paper.

. In [22,23,24], it is noted that the feed structure of the proposed pyramid horn antenna accounts for a significant portion of its overall volume. However, the feed structure of the pyramidal horn antenna designed in this paper accounts for a minimal proportion of the entire antenna structure, thereby enabling the majority of the antenna structure to be effectively utilized for the radiation of electromagnetic energy. Meanwhile, the compact feed structure enhances the integration of the antenna with the radar chip. Additionally, the bandwidth of the pyramid horn antenna based on coupled mode feeding proposed in [24] is extremely limited. However, while ensuring optimal antenna performance, the SIW quasi-pyramid horn antenna incorporates a more compact feed structure. Compared to the millimeter wave metal horn antenna proposed in [18], the radiation efficiency of the antenna array proposed in this paper is nearly equivalent to its radiation efficiency, while being easy to machine and cost-effective.

4.2. Error Analysis

To further assess the impact of errors on the performance of the proposed antenna array, an antenna structure model with angle errors and displacement errors was established. Among them, the angle error consists of rotation error and tilt error, while the displacement error consists of gap error and translation error. The antenna structure model with angle error is illustrated in Figure 20a, while the antenna structure model with displacement error is illustrated in Figure 20b,c. The antenna structure model is in its initial position of the structural model as shown in Figure 15 when no angle error and displacement error are applied. The rotation error occurs when the dielectric substrate rotates at a certain angle around the z-axis in the xyz coordinate system, while the tilt error occurs when the dielectric substrate tilts at a certain angle around the y′-axis in the x′y′z′ coordinate system. In Figure 20b, the gaps between each layer of dielectric substrates are designated as g. In Figure 20c, the offset distances of the dielectric substrate along the x and y directions are specified as Px and Py, respectively. The influence of angular errors (comprising rotation errors and tilt errors) on antenna performance is illustrated in Figure 21a,b. The angles of rotation and tilt of the dielectric substrate were set to 0.2°, 0.4°, and 0.6°, respectively. The influence of gap error on antenna performance is illustrated in Figure 21c,d. The gaps between each layer of dielectric substrates are set at 0.01 mm, 0.02 mm, and 0.03 mm, respectively. The influence of translation error on antenna performance is illustrated in Figure 21e,f. The translation distances of the dielectric substrate along the x-direction and y-direction are set at 0.02 mm, 0.04 mm, and 0.06 mm, respectively. It can be seen from Figure 21 that the gap error has a greater influence on antenna performance. Angular errors and translation errors have a minimal impact on antenna performance. Furthermore, it is crucial to minimize the gap error between each layer of the dielectric substrate in order to ensure optimal antenna performance. In Figure 21d, when the gap between each dielectric substrate is set to 0.03 mm, the realized gain of the antenna array experiences a significant reduction, and the radiation pattern in the xoz plane and yoz plane also undergoes significant distortion. Therefore, in the practical application of the proposed antenna, it should be possible to ensure a tight fit between the dielectric substrates of each layer.

5. Conclusions

This paper presents an SIW quasi-pyramidal horn antenna based on patch coupling feed that is easy to machine and integrate with radar chips. Compared to the metal pyramid horn antenna, the stepped SIW quasi-pyramid horn antenna is easier to machine, has a lower cost, and exhibits improved tolerance for machining errors. The patch coupling feed structure offers a solution for integrating the horn antenna and the radar chip. Additionally, it effectively reduces the size of the feed structure. The antenna element is capable of operating within the frequency range of 74.1 GHz to 82.3 GHz (covering the 76–81 GHz working frequency band of the automotive radar), with a realized gain of 9 dBi and a radiation efficiency of 95%. The antenna array is capable of operating within the frequency range of 74.3 GHz to 81.52 GHz (covering the 76–81 GHz working frequency band of the automotive radar), with a realized gain of 14.56 dBi and a radiation efficiency of 80.9%. Error analysis reveals that the proposed antenna exhibits strong compatibility with machining errors. These factors indicate that the proposed antenna element can be utilized as an array element for both conventional arrays and MIMO arrays applied to automotive radar sensors.