Ka-Band Wide-Angle Scanning Phased Array with Dual Circular Polarization

Abstract

A wide-angle scanning phased array with dual circular polarization in the Ka-band is presented in this paper. To improve the scanning capability of the phased array, the microstrip element is modified by loading many metal posts at its center and periphery. In addition, a stripline coupler is designed to achieve dual circularly polarized (CP) radiation, and the inner conductor of the subminiature micro-push-on (SSMP) connectors for feeding the coupler is extended to the top layer of the multilayer element by introducing an open stub, which simplifies the assembly process between the SSMP connector and multilayer printed circuit board (PCB) due to through-hole soldering instead of blind-hole soldering. The proposed element can cover a frequency range from 28 GHz to 30.5 GHz with a relative bandwidth of 8.5% in the Ka-band. An 8 × 8 phased array is constructed based on this proposed element, and a wide-angle scanning range from −65° to +65° is obtained for the dual circular polarization. The proposed array has a gain fluctuation of 5.1 dB and an axial ratio (AR) of less than 3.3 dB during beam-steering. The prototype is fabricated and measured, with a good agreement between the measured and simulated results. The proposed phased array can be applied in a Ka-band millimeter-wave (MMW) communication system.

1. Introduction

Recently, with the increasing demand for higher data rates and wider available bandwidth, communication in the MMW band has attracted much attention [1,2,3]. Although sufficient spectrum resources can be obtained in the MMW band, the terminal equipment for MMW communication also suffers from high path loss [4]. In addition, it also needs to achieve coverage over a wide scanning range in order to provide uninterruptible communication when integrated on the move [5,6,7,8]. As a result, a phased array with wide-angle scanning capability is preferred due to its high gain and agile beam-switching [9,10,11]. Compared with linear polarization, the CP phased array has the advantages of anti-polarization interference, anti-multipath fading, and immunizing Faraday rotation, which makes it widely used in satellite communication systems [4]. Therefore, the investigation of CP phased arrays with wide-angle scanning capability in MMW is of great significance [12].

Many CP phased arrays with beam-steering capability in the MMW band have been reported [13,14]. The most commonly used method to achieve CP radiation is to chamfer the patch of the microstrip antenna [15]. For example, ref. [13] presents a wideband CP array covering K/Ka-band. By truncating the corner of the stacked patch antenna and using the sequential rotation technique, a good left-hand circular polarization (LHCP) quality can be achieved. The array can scan at an angle of ±38° with a gain drop of 3 dB. By feeding the patch antenna through an asymmetric aperture, CP radiation can also be achieved by degenerating two orthogonal modes with equal amplitudes and a 90° phase difference [16,17]. In [16], a 1 × 4 passive CP phased array in Ka-band is presented based on an L-shaped slot-coupled CP antenna element. The array has a scan angle from 0° to ±38° with a gain loss of 1 dB and an AR of less than 3 dB. In addition to the microstrip antenna, many other types of antennas can be used to design the CP phased array such as dipoles [18], magneto-electric dipole antennas [19], endfire antennas [20] and so on. For example, a 16 × 16 Ku-band CP phased array is proposed in [18], based on a cross dipole. By introducing a metal cavity, the mutual coupling between the elements is reduced and the scanning capability of the proposed array can be improved. The array can achieve beam-scanning of ±40° within a bandwidth of 10.1%. The gain fluctuation is less than 3 dB, and the AR is less than 2 dB during scanning. In addition to the CP element, the phased array can also achieve CP radiation based on the linearly polarized (LP) element by employing a polarization converter [14,21] and sequential rotation technology [22]. For example, in [14], a Ka-band LP substrate integrated waveguide (SIW) slot array can be converted into a CP antenna by introducing a hybrid polarizer. The proposed array has a maximum scanning angle of 60° with an AR of less than 3 dB.

The majority of the research on CP phased arrays in the MMW band focus on single circular polarization, and only a few studies on dual-CP phased arrays in MMW band have been reported [23,24,25,26]. However, the dual-CP phased array can increase channel capacity compared to the single CP phased array. Hence, it has a better perspective. A simple method to achieve dual-CP radiation is to excite a single CP element by two orthogonal ports. For example, in [23], the stacked square patch antenna is chamfered and fed by two orthogonal ports to achieve dual-CP radiation. Here, the phased array can achieve beam-scanning at angles of ±60° with a scan loss of 4.6 dB, while the maximum AR, at an angle of 60°, is less than 5 dB. In [24], a stacked CP element with a chamfer is orthogonally fed by two slots to achieve dual-CP radiation. Meanwhile, sequentially rotated technology is employed to further improve the AR. The array can scan the main lobe at angles of ±60° with a gain loss of less than 4.5 dB. At the boresight, the AR of the array is less than 2 dB, while this increases to 12 dB when the beam is scanned at a large angle. Compared to the single-fed dual-CP element mentioned above, by exciting the element with the equal amplitude and orthogonal phase, dual-CP radiation can be achieved in a wider bandwidth and a better AR can be obtained. There are two common methods to achieve this. One is to directly excite the element by a dual TR channel. As shown in [25], the dual-LP aperture-coupled array is fed by the silicon beamformer chips and dual-CP radiation can be achieved. The array has a scan angle of ±50° with a scan loss of less than 3 dB and an AR of less than 3 dB. The other method is to excite the dual-LP element with a 3 dB-coupler. In [26], a dual-CP SIW cavity-backed antenna is proposed. By employing an SIW directional coupler, dual-CP radiation can be generated. The proposed dual-CP array, with eight elements, can steer the main beam from −59° to +58° with a gain loss of less than 3.2 dB. With the development of digital signal-processing technologies, digital beamforming techniques are applied to realize flexible beam control without employing expensive phase-shifting circuits or components [27,28]. In [27], a digital beamforming array in K-band is proposed. The digital beamforming array has a 2-D beam scanning capability and can achieve beam-scanning within ±40° with a gain fluctuation of less than 3 dB. In [28], an eight-element dipole array is proposed. Due to the optimization of the phase delay factors, the proposed array can steer its beam to any direction with high directionality and low complexity. Unfortunately, only the LP antennas are employed in [27,28]. From the previous research mentioned above, the dual-CP phased arrays in the millimeter-wave bandwidth still suffer from a limited scanning range and worse AR when beam-steering in a wide range, and their scanning capability should be further improved.

In this paper, a Ka-band dual-CP phased array with a wide-angle scanning capability is proposed for MMW communication. The conventional stacked microstrip element is used as the unit cell and is modified to improve the scanning capability by surrounding metal posts in the inner and at the periphery of the unit cell. A modified 3 dB strip-line coupler is introduced to achieve dual-CP radiation. To simplify the assembly process, the inner conductor of the SSMP connector is extended to the top layer of the multi-layer PCB, which can replace a complex operation of the blind-hole soldering between the SSMP connector and the multi-layer PCB employed in [23] with a simple operation of the through-hole soldering. An open stub is used to compensate for the coupler’s reflection coefficient deterioration due to the extended inner conductor. Based on this unit cell, an 8 × 8 phased array is designed, simulated, fabricated, and measured, and the measured results show reasonable agreement with the simulated ones.

The paper is organized as follows. In Section 2, the proposed unit cell in an infinite array is designed and analyzed, including the dual-polarized wide-angle scanning antenna element and the modified strip-line coupler. Based on the proposed unit cell, a finite phased array with 8 × 8-unit cells is constructed to achieve the wide-angle scanning capability for dual polarization. The finite array is simulated, fabricated, and measured, and the results are discussed in Section 3. Finally, a brief conclusion is given in Section 4.

2. Design of Unit Cell in Infinite Array

2.1. Configuration of the Proposed Unit Cell

The configuration of the proposed unit cell is shown in Figure 1. It consists of a stacked antenna, a strip-line coupler, the shielding posts, vertical RF transition and two SSMP ports for feeding, as shown in Figure 1a. The stacked antenna is employed here to obtain a wide bandwidth. It is composed of a driven patch in Layer 2 and a parasitic patch in Layer 1, as depicted in Figure 1c,d. The patches are cut in the corner and center to provide more degrees of freedom when optimizing the performance. The strip-line coupler is below the stacked antenna and feeds the antenna with equal amplitude and orthogonal phase to achieve dual-CP radiation. The strip-line coupler is the ring structure, shown in Figure 1f. For easier fabrication and higher cost efficiency, the strip-line coupler is directly connected to the driven patch by the vertical RF transition instead of coupling by the slots. The stacked antenna and the strip-line coupler are integrated by employing the multi-layer PCB process and consist of three substrate layers bonded by two prepreg layers, as shown in Figure 1b. The metal shielding posts are introduced and distributed in the internal and at the periphery of the unit cell. For the antenna, these posts can suppress the surface wave effect, which contributes to enlarging the scanning coverage and improving the isolation between two ports [29]. For the strip-line coupler, these posts can be used to prevent the parallel-plate mode [30]. Two SSMP connectors are used to excite the unit cell. As usual, the inner conductor of the SSMP connector ends at the layer where the coupler is located, and blind-hole soldering is needed when assembling the SSMP connectors onto the multi-layer PCB. For simplifying the assembly process, the inner conductor of the SSMP connector is extended to the top layer of the multi-layer PCB, which can eliminate the requirement for blind-hole soldering. Four metal posts are distributed around the inner conductor of the SSMP connector to form a coaxial-like structure. Furthermore, two ground planes are added in Layer 1 and Layer 2, as shown in Figure 1c,d. The inner conductor of the SSMP connectors passes through the ground plane of Layer 2 without contact and is directly connected to the ground plane of Layer 1. This short-circuited design purposely facilitates the welding of the SSMP connectors. For mitigating the negative influence of the short-circuit design on the coupler’s performance, an open stub is introduced, as shown in Figure 1f. The unit cell covers a range from 28 GHz to 30.5 GHz, and its size of L₁ is set to 5 mm to meet the inter-element spacing requirement when the main beam-steering is conducted at angles of ±65° without grating the lobe at 30.5 GHz. The proposed unit cell is simulated by a high-frequency structure simulator (HFSS) with master/slave boundaries. The optimized parameters are listed in

Table 1. Structure parameters of the proposed unit cell (unit: mm).

Param.	Value	Param.	Value	Param.	Value
h0	0.1	d1	1	L1	5
h1	0.762	d2	0.8	L2	0.3
h2	0.254	d3	1.05	L3	2.17
h3	0.127	d4	0.71	L4	3.2
w1	0.28	d5	2	L5	0.85
w2	0.16	d6	0.6	L6	2.5
w3	0.5	d7	1.1	L7	0.4
w4	0.3	d8	2.2	r5	0.45
r1	0.86	r3	0.3	r6	0.6
r2	0.92	r4	0.3

.

2.2. Design of the Antenna Element in the Unit Cell

First, a traditional stacked microstrip antenna element without metal posts is employed as the original element, as shown in Figure 2a. To evaluate the performance of the element, the original element is directly excited by two SSMP connectors instead of a strip-line coupler. The parameter names in Figure 1 are fully applicable to Figure 2 and they will not be repeated in Figure 2, for clarity. The original stacked element is composed of a driven patch and a parasitic patch, in which the driven patch is located on the lower substrate with a thickness of h₂ and the parasitic patch is mounted on the upper substrate with a thickness of h₁; these two substrates are bonded by a prepreg layer with a thick of 3 × h₀. The driven patch and the parasitic patch are centered vertically. By adjusting the size (L₃ and L₆) of two patches, two resonant frequencies can be close to each other, and a wide bandwidth can be obtained. Two SSMP connectors are arranged along the diagonal direction to excite the element. By optimizing the position (L₅) of the SSMP connectors, the impedance matching of the element can be adjusted. In addition, some defective structures are introduced to help tune the element more flexibly, including two sectors (L₇), a circle (d₁) and four squares (L₂) distributed at the edge and the center of the patches when optimizing by HFSS.

To investigate its performance at different scanning angles, the original element was simulated in the master/slave boundary. The simulated results, including the active voltage standing wave ratio (VSWR) and isolation, are shown in Figure 3. Owing to the symmetric structure of the original element, only the active VSWR of port 1 is illustrated here. It can be seen that, when the main beam-scanning in the angle range of 45°, the active VSWR is less than 2 dB. Meanwhile, the isolation between two ports is better than −10.5 dB in the desired bandwidth of 28 GHz to 30.5 GHz. However, when the main beam-scanning to 60° and 65°, the maximum active VSWR increases to 3.5 and 4.2, respectively. What is more, the isolation deteriorates to −3.8 dB and −3.7 dB. The electric field distribution at the center frequency of 29.25 GHz is depicted in Figure 2b,c when beam-scanning at 45° and 65°, respectively. Since the original element is fed by port 1 along the diagonal direction, the electric field should be distributed along the diagonal direction, as shown in Figure 2b, which means that a good active VSWR and isolation can be obtained when beam-scanning at 45°. When beam-scanning at 65°, the electric field shown in Figure 2c has been changed and distributed along thex-axial direction, which will lead to a large mutual coupling between two ports. In conclusion, the original element has a limited scanning coverage of ±45° and some measures should be taken to enlarge the scanning range.

To improve the scanning ability, the original element was modified and named as modified element, shown in Figure 2d. The GND planes were introduced in the Layer 1 and Layer 2, and many metal posts with a diameter of r₅ and distance of d₃ were loaded and distributed at the periphery of the modified element. As a result, an SIW cavity with a size of L₄ × L₄ × (3 × h₀ + h₁ + h₂) was established. These loaded posts can suppress the surface wave effect and re-distribute the electric field [29]. As shown in Figure 2e,f, by introducing the metal posts, the electric field of the modified element is re-distributed along the diagonal direction when beam-scanning at 65°, which is consistent with the direction of port 1. The isolation between two ports can also be improved. The simulated results are shown in Figure 4. It can be seen that the isolation of the modified element between two ports can be optimized to less than −10.5 dB when beam-steering at an angle of 65° in the xoz and yoz plane, as shown in Figure 4b,d. Furthermore, the modified element has a better active VSWR of less than 2 when beam-scanning in the xoz or yoz plane, as shown in Figure 4a,c. The simulated results indicate the modified element has a wide-angle scanning ability in the desired bandwidth from 28 GHz to 30.5 GHz within a fractional bandwidth of 8.5%.

2.3. Design of the Strip-Line Coupler

To achieve dual-CP radiation, a modified strip-line coupler with equal amplitude and orthogonal phase was designed. Figure 5 shows the evolution of the strip-line coupler. Firstly, an original strip-line coupler is designed in Figure 5a, and its stack-up is shown in Figure 5d. The original coupler is a ring structure with four ports, in which port 1 and port 2 are input ports for right-hand circular polarization (RHCP) and LHCP, respectively, and port 3 and port 4 are output ports to achieve power divider with equal amplitude and 90° phase difference. Two SSMP connectors are connected with port 1 and port 2, and their inner conductors end at the layer 4 where the coupler is located. Figure 6a shows the simulated performance of the original coupler. It can be seen that the reflection coefficient of input ports is less than −21.9 dB, while the isolation between them is less than −20.9 dB in the desired bandwidth from 28 GHz to 30.5 GHz. Furthermore, a good amplitude consistency of ±0.1 dB and phase difference of 90 ± 1° between the output port 3 and port 4 can be obtained. The simulated results indicate that the original coupler can be used to excite the modified antenna element in Section 2.2 for dual-CP radiation. Then, the substrates of the modified antenna element were integrated with the strip-coupler, shown in Figure 5b. To avoid blind-hole soldering when assembling the SSMP connectors onto the multi-PCB, the inner conductors of the SSMP connectors were extended to the top Layer 1 and short-circuited with the ground in Layer 1, as in Figure 5b,e. This operation will facilitate the welding of SSMP connectors. However, this design is equivalent to the parallel of a short-circuiting stub for input ports and will deteriorate the performance of the coupler. As shown in Figure 6b, the reflection coefficient of the input port was deteriorated and maintained between −10 dB and −12 dB. In order to address the issue above, an open stub is introduced to the input ports shown in Figure 5c. By optimizing the parameters of the stub, a good S₁₁ lower than −18.6 dB and isolation between input ports of less than −21.3 dB was attained, as shown in Figure 6c. In addition, the output ports have a good amplitude consistency of ±0.18 dB and phase difference of 90 ± 1.1° and can be used to excite the modified antenna element for a dual-CP radiation.

2.4. Performance of the Unit Cell in Infinite Array

The modified antenna element proposed in Section 2.2 and the modified strip-line coupler 2 proposed in Section 2.3 were integrated into the dual-CP unit cell shown in Figure 7. It was simulated in a periodic boundary to investigate its scanning performance in the xoz and yoz plane. Owing to the symmetric structure of the proposed unit cell, only the performance of port 1 is illustrated in Figure 8. It can be seen that the proposed unit cell has a good active VSWR in the frequency range of 28~30.5 GHz, no matter when beam-scanning is conducted along the xoz plane and yoz plane. In the xoz plane, the active VSWR is less than 2.2 within a scanning angle of 65°, while that is less than 3 in the yoz plane. The isolation between two input ports was kept below −11.8 dB and −12.1 dB, respectively. The simulated results demonstrate that the proposed unit cell has a good performance at a scanning angle of ±65° and can be used to construct a dual-CP phased array with wide-angle scanning ability.

3. Finite Array Simulation and Measurement

3.1. Finite Array Design and Simulation

A finite phased array based on the proposed unit cell was designed to achieve wide-angle scanning at ±65° in a bandwidth ranging between 28 GHz to 30.5 GHz. The phased array consists of 8 × 8-unit cells arranged in a rectangle lattice; its model for the simulation is shown in Figure 9. The inter-element spacing between the unit cell is 5 mm to ensure no grating lobe when beam-steering at the angle of ±65° at 30.5 GHz. To simulate the situation in a practical application so that each unit cell is directly excited with different amplitudes and phases by TR channels to achieve beam-steering in a wide-angle range, the proposed phased array is individually fed by 128 ports to each unit cell rather than two 8 × 8 networks for LHCP and RHCP array because a well-designed network for a certain scanning angle cannot be applicable to beam-forming for other scanning angles. In this simulation, by setting the amplitude and phase of each unit cell’s excitation, the scanning performances of the proposed phased array in a wide scanning angle can be obtained. The finite phased array was simulated at 28 GHz, 29.25 GHz and 30.5 GHz, respectively. Owing to the symmetric structure, there is a similar radiation performance for LCHP and RHCP. As a result, only LHCP radiation performance is exhibited here for brevity in conditions such that the LCHP ports are excited, while RHCP ports are not excited, and are terminated by a lumped port with 50 Ω port impedance in the simulation.

Figure 10 shows the radiation pattern when beam-steering occurs along the xoz-plane and yoz-plane, respectively. It can be seen that the phased array can steer its main beam in the angle range of ±65° in the desired bandwidth of 28 GHz to 30.5 GHz. The simulated scanning gain and corresponding AR are summarized in

Table 2. Simulated performances of the proposed phased array.

Scanning in xoz Plane		−65°	−45°	−30°	−15°	0°
28 GHz	Gain(dB)	18	20.7	21.6	22.1	22.2
	AR(dB)	1.4	0.4	0.6	0.9	0.9
29.25 GHz	Gain(dB)	18.4	21.1	22	22.4	22.6
	AR(dB)	1.7	0.4	0.1	0.2	0.2
30.5 GHz	Gain(dB)	17.9	21.5	22.4	22.9	23
	AR(dB)	0.9	0.3	0.3	0.5	0.4
Scanning in yoz Plane		−65°	−45°	−30°	−15°	0°
28 GHz	Gain(dB)	18.2	20.8	21.6	21.2	22.2
	AR(dB)	3.1	1.8	1.3	1	0.9
29.25 GHz	Gain(dB)	17.9	20.9	21.9	22.4	22.6
	AR(dB)	3.3	1.6	2.5	0.4	0.2
30.5 GHz	Gain(dB)	17.9	21.3	22.3	22.8	23
	AR(dB)	1.8	1.8	1.1	0.6	0.4

. Benefiting from the modified coupler, a good AR versus scanning angle can be observed. In the scanning range of 65°, the simulated AR is kept less than 3.1 dB, 3.3 dB and 1.8 dB at 28 GHz, 29.25 GHz, and 30.5 GHz, respectively. Furthermore, the scanning gain is maintained between 18 dB and 22.2 dB, 17.9 dB and 22.6 dB, 17.9 dB and 23 dB, indicating a maximum scan loss of 4.2 dB, 4.7 dB and 5.1 dB at 28 GHz, 29.25 GHz, and 30.5 GHz, respectively.

3.2. Fabrication, Measurement and Discussion of the Finite Phased Array

In order to verify the design, the proposed finite phased array was fabricated and measured; its prototype is shown in Figure 11. The SSMP connectors are employed to excite the unit cells of the phased array. Benefiting from the extended and short-circuit design of the inner conductor, the SSMP connector can be easily assembled and welded by a through-hole soldering process. Due to the large size of the adopted SSMP connector, it is difficult to arrange two SSMP connectors in a unit cell for dual-circular polarization. As a result, the ports for LHCP are connected with the SSMP connectors, while the ports for RHCP are kept in an open-circuit state. In practical applications, LHCP and RHCP ports can be connected to the TR channel by customized compact coaxial insulators or directly integrated with a beamformer chip with a multi-layer PCB process as introduced in [24]; thus, the SSMP installation problem for dual polarization can be avoided. In order to compare the simulated and measured results more rationally, the phased array shown in Figure 9 was re-simulated in a case where the LHCP ports were excited and the RHCP ports were kept in an open-circuit state.

The S-parameters of the embedded element in the phased array were measured. Unfortunately, the isolation between RHCP port and LHCP port in a unit cell cannot be measured due to the open-circuit states for RHCP ports. Since there is a difference in performance between elements at the edge and those in the center, the four elements shown in Figure 11 were selected and their S-parameters were simulated and measured, as shown in Figure 12. It can be seen that in the desired bandwidth ranging between 28 GHz to 30.5 GHz, the simulated reflection coefficient is less than −15.7 dB and −13.1 dB for S₁₁ and S₃₃, while the measured one is less than −16.2 dB and −13.5 dB, respectively. The isolation between adjacent elements was measured, as shown in Figure 12b,d. The simulated isolation was better than −15.8 dB and −13.8 dB for S₁₂ and S₃₄ in the desired bandwidth, while the measured one was better than −12.2 dB and −15.5 dB, respectively.

Although the 8 × 8 feed network with power dividers and phase shifters can be used to measure the scanning performances of the proposed array, it is a complex and expensive operation because a well-designed feed network only works well for a certain beam and a certain frequency point, and a variety of networks with different phased shifters are required for the pattern measurements of a multi-frequency point and multi-scanning beam. As a result, a more simple and cost-efficient method based on the measured active element patterns was employed in this paper. The scanning performance of the proposed phased array can be calculated from the measured active element patterns (AEPs) [31]. When one element is measured, the other elements are terminated with matching loads. The scanning radiation pattern of the phased array can be calculated as the superposition of AEPs of all elements, as follows:

$$F_{array}={\displaystyle \sum }_{n=1}^8{W}_n{f}_n\left(\theta \right)e^{j\left(k\overrightarrow{r}·r_n-{\varnothing }_n\right)}$$

(1)

where f_n is the AEP of the nth element, and W_n and ${\varnothing }_n$ are the feeding amplitude and the phase of the nth element, respectively, r_n and r are the position vectors for the origin to the center of the nth element and the unit vector of the beam pointing direction, respectively. In this experiment, the W_n is set to 1 because no amplitude tapering is applied.

The measured and simulated radiation patterns along the xoz-plane and yoz-plane at 28 GHz, 29.25 GHz and 30.5 GHz are shown in Figure 13. As can be seen, the phased array can steer its main lobe at an angle of 65°, indicating a wide-angle scanning ability. The simulated and measured gain and AR versus the scanning angle are depicted in Figure 14. The simulated gain is maintained between 22.3 dB and 17.8 dB at 28 GHz, 22.6 dB and 17.8 dB at 29.25 GHz, 22.9 dB and 17.7 dB at 30.5 GHz, respectively, and the simulated gain loss of 4.5 dB, 4.8 dB and 5.2 dB can be obtained. The simulated scanning AR is less than 4.9 dB, 5 dB and 3.9 dB, respectively. Compared with the results shown in Table 2 with RHCP ports terminated by a 50 Ω load, the open-circuit state for RHCP ports has an impact on the phased array’s performance, especially on the AR value, which is caused by the imperfect isolation between LHCP port and RHCP port in a unit cell; this is the reason why the phased array was re-simulated to keep it consistent with the fabricated prototype. The measured gain was maintained between 21.7 dB and 17.5 dB at 28 GHz, 22.5 dB and 18.3 dB at 29.25 GHz, 22.4 dB and 16.9 dB at 30.5 GHz, respectively, and a gain loss of 4.2 dB, 4.2 dB and 5.5 dB can be obtained. The measured scanning AR was less than 4 dB, 4.2 dB and 3.2 dB, respectively. The measured results agreed with the simulated ones, which validated the design.

Table 3. Comparison of dual-CP phased array in millimeter-wave band.

	[24]	[23]	[25]	This Work
Frequency	29.5~30 GHz	18.8~20 GHz	17.7~21.2 GHz	28~30.5 GHz
Bandwidth	1.7%	6.2%	18%	8.5%
No. of elements	32 × 32	8 × 8	8 × 8	8 × 8
Scan Range	±60°	±60°	±50°	±65°
Scan loss	4.5 dB	4.6 dB	3 dB	5.1 dB
AR	<12 dB	<5 dB	<3 dB	<3.3 dB
Isolation	NA	<−15 dB	<−10 dB	<−12 dB
Complexity	moderate	moderate	moderate	moderate

shows a comparison between this work and other reported dual-CP phased arrays with 2D wide-angle scanning ability in a millimeter-wave band in terms of performance and complexity. In terms of performance, Refs. [23,24] aimed to achieve the dual-CP radiation based on the single-fed patch with a corner cut. As a result, they have a narrow AR beamwidth in nature and the AR will deteriorate when beam-steering at a large angle. Ref. [25] is based on silicon beam-former chips used to excite the dual-LP ports with equal amplitude and an orthogonal phase to achieve the dual-CP radiation, and a good AR can be obtained when beam-scanning. However, the scanning range of [25] is limited to 50° owing to deterioration of the isolation and active VSWR. In this work, compared with the other reported phased arrays listed in Table 3, a wider scanning range and a better AR were observed. Although the isolation between two ports for dual polarization is not perfect as [23], it has been kept below −12 dB in the whole band when main beam-steering is conducted at an angle of ±65°. In terms of complexity, a multilayer PCB stackup of more than 10 layers was employed in [24,25] to achieve the integration of the phased array and the beamformer chips, indicating a potentially more complex structure. However, neglecting the integrated TR channels and just considering the antenna, including network and radiation aperture, this work has a similar multilayer PCB stackup to [23,24,25]. Compared with [23,24,25], this work has a simpler assembly process without the blind-hole soldering.

4. Conclusions

In this paper, a Ka-band dual-CP microstrip element was designed that ranged between 28~30.5 GHz with a relative bandwidth of 8.5%. In order to improve the scanning capability, metal posts were inserted at the center and periphery of the antenna element to suppress the surface wave and re-distribute the electric field. As a result, the isolation between the ports was improved and the scanning range could be extended. To achieve dual-CP radiation, a modified strip-line coupler with an open stub was designed and integrated into the antenna element. For ease of assembly, the inner conductors of the SSMP connectors were extended to the top layer and short-circuited to the ground of the multi-layer PCB, thus eliminating the need for blind-hole soldering. The proposed element has a good active VSWR of less than 3 and isolation better than −11.8 dB at a scanning angle of ±65°. An 8 × 8 phased array was established based on this element. The array can scan the main beam at an angle range of ±65°, where the gain loss is less than 5.1 dB, and the AR is less than 3.3 dB. A prototype was fabricated and measured, and the measured results are in agreement with the simulated ones. The proposed phased array has the features of wide-angle scanning and dual-CP radiation, and is a good candidate for Ka-band MMW communication systems.