An Active Clamp Dual-Inductor Isolated Current Source to Current Source Converter with Wide Output Voltage Range

Abstract

Human observation of the ocean has gradually evolved from the sea surface to systematic monitoring and sampling through seafloor observation networks, and constant current power supply has become the main power supply method for seafloor observation networks due to its high reliability. There are some studies on current source to voltage source converters, but there are few studies on current source to current source (CS/CS) converters, which affects the expansion of power supply networks for seafloor observation networks. In this paper, by employing input current sharing and output voltage doubling circuits, an active clamp dual-inductor isolated CS/CS converter which uses a single-stage conversion circuit to realize constant current source conversion with a wide output voltage range is proposed. Active clamp technology at the primary side of the proposed circuit is employed to recover energy stored in leakage inductance, suppress voltage spikes of the primary side switches, and achieve zero-voltage switching of the primary side switches. The secondary side’s output voltage doubling circuit resonates with transformer leakage inductance to achieve zero-current switching of the secondary side diodes, which can reduce losses and enhance efficiency. The operating principles of the proposed circuit are analyzed in detail, and the characteristic and parameter design analysis, including current conversion ratio, transformer turn ratio, power inductors, and resonant capacitors and inductor, are presented. Finally, the experimental results based on a 100 W experimental prototype validate the feasibility of the proposed converter.

1. Introduction

Oceans play a crucial role in maintaining biodiversity, providing economic resources, and human evolution. The vast waters of the ocean provide a regulatory role for the global climate system. At the same time, marine organisms in marine ecosystems participate in complex ecological cycles, affecting key ecological processes such as the global carbon and oxygen cycles. In addition, the oceans are rich in mineral and biological resources, such as oil, natural gas, metallic minerals, and fish. These resources are of great significance for human economic development and social progress. Therefore, ocean observation techniques and equipment are also constantly evolving. Observation methods for the ocean have extended from traditional surface observation to seabed observation [1,2,3,4,5]. As early as the early 1970s, countries around the world had begun researching underwater observation networks. Some of the well-known ocean observation systems around the world include the Martha’s Vineyard Coastal Observatory in Massachusetts, USA, the Hawaii-2 Observatory in Hawaii, the Monterey Accelerated Research System in California, the North-East Pacific Time-series Undersea Networked Experiments in Canada, the Advanced Real-Time Earth Monitoring Network in the Area in Japan, and the development of the Dense Ocean-floor Network System for Earthquakes and Tsunamis [6,7,8,9,10].

The power supply method with a direct current (DC) input constant current source exhibits high reliability and high efficiency, which is crucial for the long-term stable operation of underwater observation equipment [11]. In addition, even if a short circuit fault occurs in seawater, a constant current power supply can also supply the current on the fault-free cables to the load next to the fault point through an energy splitter, achieving continuous power supply to the system [12,13,14]. Therefore, constant current power supply is a reasonable and widely used method for the power supply of submarine observation networks with high reliability requirements [15].

A block diagram of the power supply structure of the common submarine observation networks powered by a constant current source is shown as Figure 1. The submarine observation network has a large span and is mainly composed of power feeding equipment (PFE), power branching units (PBUs), and observation nodes [16]. Among these, the shore-based high-voltage power supply equipment PFE outputs constant current electrical energy to underwater equipment. Usually, the power supply in an observation node which is connected in series with the PFE uses a current source to voltage source converter to provide electrical energy for the equipment in the observation node. Each PBU receives electrical energy from the PFE and extends electrical energy to other observation nodes which are not connected to the PFE. Due to the use of a constant current power supply in the seabed observation network, each PBU needs use a current source to current source (CS/CS) converter to provide electrical energy to the observation nodes which are not connected to the PFE; therefore, research on CS/CS converters is very necessary [17,18,19].

Currently, traditional designs for converters with current source inputs employ a Zener diode matrix to convert a constant current source into a relatively stable voltage source [20]. Then, this voltage source which is generated by Zener diodes is transformed to the desired voltage for the observation equipment through an isolated DC–DC converter. When the current source flows through the Zener diode matrix, and the DC–DC converter in parallel with the Zener diode matrix, the smaller the output power of the converter, the greater the power loss of the Zener diodes, which will lead to very low efficiency and high losses with light loads. When the output load is light, it is difficult to dissipate the power consumed by the Zener diodes through natural heat dissipation, which will lead to thermal issues and reliability issues.

In [21], a basic circuit of a push–pull converter with an input inductor is proposed to achieve current source to voltage source conversion. However, this solution suffers from voltage spikes because of the leakage inductance of the transformer, and a specialized start-up and auxiliary power supply circuit need to be designed for the method. In [22], some scholars proposed a current source input converter based on full bridge resonance which adopts a single-stage circuit structure and can achieve a larger power output. The full bridge switch can achieve zero-voltage turn-on to reduce the turn-on loss of the switch and improve the efficiency of the converter. However, this converter in [22] is difficult to adapt to light-load operation and requires setting a minimum load. In [23], by using a current-fed three-phase three-leg bridge, three single-phase rectifier bridges, and three single-phase transformers with a primary star connection, a new three-phase current-fed isolated converter is presented to achieve a constant current source input, high gain, and medium power level power source. Considering its multiphase feeding design, synchronization and coordination control between different phases are crucial, and accurate control logic timing is required to adjust each switch to ensure a stable output voltage. In [24], by cascading a pre-stage push–pull current-fed circuit and a post-stage boost constant voltage converter, a two-stage isolated current source to voltage source converter for cabled underwater information networks was proposed. However, the topology uses a two-stage converter, which will affect the efficiency and power density. The topologies in [21,22,23,24] feature converters which convert current source to voltage source, but these converters cannot be used in PBUs because they cannot provide constant current source output. In [5,25], the scholars used push–pull topology to achieve a CS/CS converter where the output current’s average magnitude is proportional to the input current’s average magnitude, but the output current value cannot be regulated to a constant value because of the absence of closed-loop control. This converter can realize an output current source with a wide output range, but the efficiency of the converter is not high, especially with low output power, because of the hard switch.

By employing input current sharing and output voltage doubling circuits, an active clamp dual-inductor isolated CS/CS converter which uses a single-stage conversion circuit to realize constant current source conversion with a wide output voltage range is proposed in this paper. To enhance efficiency, active clamp technology at the primary side of the proposed circuit is employed to recover energy stored in leakage inductance, suppress voltage spikes of the primary side switches, and achieve zero-voltage switching of the primary side switches; the secondary side’s output voltage doubling circuit resonates with the transformer leakage inductance to achieve zero-current switching of the secondary side diodes. The operating principles and the parameter design analysis of the proposed circuit, including the power inductors, transformer, and resonant capacitors, are analyzed in detail. The experimental results based on a 100 W experimental prototype validate the feasibility of the proposed converter, and the efficiency of the converter, with rated output power, can reach 91.6%.

This paper consists of five sections, as follows. The operating principle of the proposed active clamp dual-inductor isolated CS/CS converter is analyzed in Section 2. The characteristic and parameter design analysis, including the current conversion ratio, transformer turn ratio, power inductors, and resonant capacitors and inductor, are presented in Section 3. Verification results based on a 100 W experimental prototype are given to verify the feasibility of the proposed converter in Section 4. A summary of the conclusions is provided in Section 5.

2. Operating Principle of Active Clamp Dual-Inductor Isolated CS/CS Converter

2.1. Operating Principle of Main Circuit Topology and Control Circuit

The main circuit and control circuit block diagram of the proposed active clamped dual-inductor isolated CS/CS converter with a wide output voltage range is shown in Figure 2.

As shown in Figure 2a, the proposed circuit consists of the input filter capacitor C_f, input filter inductor L_f, power inductors L₁ and L₂, clamp capacitor C_c, power switches S₁, S₂, S₃, and S₄, transformer T, secondary rectifier diodes D₁ and D₂, resonant capacitors C_r1 and C_r2, resonant inductor L_lk, which is the secondary leakage inductance of the transformer T, and the output filter capacitor C_o. The active clamp circuit is employed on the primary side of the converter to suppress voltage spikes and return the energy stored in the leakage inductance of the transformer. The secondary output voltage doubling circuits can expand the output voltage range; as the capacitors of the voltage doubling circuit are resonant with the leakage inductance of the transformer T to improve efficiency, the proposed converter can achieve a wide output voltage range.

The control circuit block diagram of the proposed converter is shown in Figure 2b. The control circuit of the converter consists of the error amplifier EA, PI compensation circuit, comparator comp, sawtooth wave generator Sawtooh, D Flip-Flop, NOR gates NOR₁ and NOR₂, NOT gates NOT₁, NOT₂, NOT₃, NOT₄, NOT₅, and NOT₆, and AND gates AND₁, AND₂, AND₃, and AND₄. By sampling the output current and converting it into a voltage signal V_FB for feedback to the control system, V_FB is compared with the reference voltage V_ref to generate an error signal. The duty cycle signal V_g is obtained by comparing the sawtooth wave V_saw and error signal. Since the input power source is a current source, the duty cycle of the main switches S₁ and S₂ should be greater than 0.5, ensuring that the proposed converter always has a continuous current path from the current source to the ground. The PWM driving signals of the active clamp switches S₄ and S₃ are 180° out of phase from the driver signal of the main switches S₁ and S₂.

For the convenience of theoretical analysis, it is assumed that all switches, diodes, inductors, and capacitors in the topology are ideal components. There are ten operating modes within one switching cycle of the proposed converter. Since the proposed converter can operate symmetrically, only the five operation modes during the first half of the switching cycle are analyzed. Figure 3 and Figure 4 illustrate the operation modes and key waveforms of the proposed topology.

2.2. Operation Modes Analysis

Mode A (t₀ − t₁): As shown in Figure 3a, S₂ is turned on and S₁, S₃, and S₄ are turned off during Mode A. This mode represents a transient dead-time period. During this mode, the current i_L1 through inductor L₁ decreases linearly, the current i_L2 through inductor L₂ increases linearly, and the magnetic current i_Lm linearly increases. Concurrently, resonance occurs between the secondary side inductor L_lk and resonant capacitors C_r1 and C_r2, causing the secondary side diode D₁ to start conducting. Since there is an imbalance between the i_p and i_L1 currents at this point, the body diode of switch S₃ conducts. Current flows through switch S₃, resulting in zero-voltage switching for switch S₃.

Mode B (t₁ − t₂): As depicted in Figure 3b, S₁ and S₄ are turned off and S₂ and S₃ are turned on. During this mode, the power source charges the clamp capacitor C_c through L₁, i_L1 decreases linearly, and the current i_L2 through inductor L₂ increases linearly. The primary voltage v_TP of the transformer voltage is equal to the voltage v_Cc across the clamp capacitor C_c.

The secondary side inductor L_lk and resonant capacitors C_r1 and C_r2 remain in resonance. The secondary side diode D₁ conducts, causing the voltage across C_r1 to rise and the voltage across C_r2 to fall. Energy is transferred from the primary side to the secondary side until diode D₁ turns off in this mode.

During this interval, the voltages v_L1 and v_L2 across inductors L₁ and L₂ are shown as

$$v_{L1}=v_{\inf }-v_{Cc}=-L_1{\displaystyle \frac{\Delta i_{L1}}{\Delta t}}$$

(1)

$$v_{L2}=v_{\inf }=L_2{\displaystyle \frac{\Delta i_{L2}}{\Delta t}}$$

(2)

where v_inf is the input voltage after inductor L_f, and Δi_L1 and Δi_L2 are the variation of the inductor currents i_L1 and i_L2, respectively.

In Mode B, the currents through inductors L₁ and L₂ and the magnetic inductor L_m can be expressed as

$$i_{L1}(t)=i_{L1}(t_1)+{\displaystyle \frac{v_{\inf }-v_{Cc}}{L_1}}(t-t_1)$$

(3)

$$i_{L2}(t)=i_{L2}(t_1)+{\displaystyle \frac{v_{\inf }}{L_2}}(t-t_1)$$

(4)

$$i_{Lm}(t)=i_{Lm}(t_1)+{\displaystyle \frac{v_{Cc}}{L_m}}(t-t_1)$$

(5)

where i_L1(t₁), i_L2(t₁), and i_Lm(t₁) are the initial values of the currents through inductors L₁ and L₂ and the magnetic inductor L_m at time t₁, respectively.

In this mode, the primary side voltage v_TP of the transformer is given as

$$v_{TP}=v_{Cc}$$

(6)

The secondary side voltage v_sec of the transformer secondary side can be expressed as

$$v_{\sec }={\displaystyle \frac{v_{Cc}}{n}}$$

(7)

where n is the ratio of the turn number of primary side to the turn number of the secondary side.

When the transformer’s secondary side is in resonance, the related resonant current of the secondary side can be given as

$$L_{lk}{\displaystyle \frac{d{i}_{Lr}(t)}{dt}}=-{\displaystyle \frac{v_{Cc}}{n}}+v_{cr1}(t)$$

(8)

$$i_{lk}(t)=C_{r1}{\displaystyle \frac{d{v}_{cr1}(t)}{dt}}-C_{r2}{\displaystyle \frac{d{v}_{cr2}(t)}{dt}}$$

(9)

where v_cr1 and v_cr2 are the voltage values across resonant capacitors C_r1 and C_r2, respectively, and i_lk is the current through the leakage inductance of transformer T.

Supposing the resonant capacitances of both C_r1 and C_r2 are same, the voltage ripple of the resonant capacitor C_r1 can be expressed as

$$\Delta v_{cr}={\displaystyle \frac{1}{C_r}}{\displaystyle {\int }_{t_1}^{t_2}{i}_{cr1}(t)dt=\frac{1}{2C_r}}{\displaystyle {\int }_{t_1}^{t_2}{i}_{D1}(t)d}t={\displaystyle \frac{T_s{I}_o}{2C_r}}$$

(10)

where T_s is switching period, C_r is the capacitance value of resonant capacitors C_r1 and C_r2, and I_o is the output current of the proposed converter.

The initial resonance situation can be assumed as

$$i_{lk}(t_0)=0$$

(11)

$$V_{cr1}(t_0)={\displaystyle \frac{V_o}{2}}-{\displaystyle \frac{\Delta V_{cr}}{2}}$$

(12)

According to Figure 3, the output voltage is the sum of the voltage resonant capacitors C_r1 and C_r2, so the output voltage can be expressed as

$$V_o=V_{cr1}+V_{cr2}$$

(13)

From (8)–(13), the current through leakage inductance and the voltage values across resonant capacitors can be given as

$$i_{lk}(t)={\displaystyle \frac{V_c+n{V}_{cr1}(t_1)}{n{Z}_r}}\sin \left[{\omega }_r(t-t_1)\right]$$

(14)

$$v_{cr1}(t)={\displaystyle \frac{V_{Cc}}{n}}+(V_{cr1}(t_1)+{\displaystyle \frac{V_{Cc}}{n}})\cos \left[{\omega }_r(t-t_1)\right]$$

(15)

where ω_r is the angular resonant frequency, Z_r is the resonant impedance, and ω_r and Z_r can be given as

$${\omega }_r={\displaystyle \frac{1}{\sqrt{L_{lk}(C_{r1}+C_{r2})}}}={\displaystyle \frac{1}{\sqrt{2L_{lk}{C}_{r1}}}}$$

(16)

$$Z_r=\sqrt{{\displaystyle \frac{L_{lk}}{C_{r1}+C_{r2}}}}=\sqrt{{\displaystyle \frac{L_{lk}}{2C_r}}}$$

(17)

Mode C (t₂ − t₃): As shown in Figure 3c, the resonance of L_lk, C_r1 and C_r2 is ended at t₂, and the current through diode D₁ becomes zero at t₂. Zero-current turn-off of diode D₁ is achieved to eliminate the reverse recovery power loss. In this mode, switches S₁ and S₄ remain off while switches S₂ and S₃ remain on.

In this mode, the system provides energy from the output filter capacitor C_o to the load, maintaining a constant output current. The current through inductor i_L1 continues to decrease with the same slew rate of Mode B, while i_L2 continues to increase with the same slew rate of Mode B as well.

Therefore, the currents through inductors L₁ and L₂, and the magnetic inductor L_m in Mode C can be given as

$$i_{L1}(t)=i_{L1}(t_2)+{\displaystyle \frac{v_{\inf }-v_{Cc}}{L_1}}(t-t_2)$$

(18)

$$i_{L2}(t)=i_{L2}(t_2)+{\displaystyle \frac{v_{\inf }}{L_2}}(t-t_2)$$

(19)

$$i_{L\mathrm{m}}(t)=i_{Lm}(t_2)+{\displaystyle \frac{v_{Cc}}{L_m}}(t-t_2)$$

(20)

Mode D (t₃ − t₄): As shown in Figure 3d, this mode represents a transient dead-time interval. In this mode, switch S₂ remains on, while switches S₁, S₃, and S₄ are off. The secondary side diodes are off. Output load energy is supplied from the output filter capacitor C_o. The voltage across the primary winding of the transformer is zero, so the current i_Lm of the magnetic inductor remains almost constant. The body diode of switch S₁ conducts to provide the current difference between the magnetic inductor current i_Lm and the current i_L1 through inductor L₁. The voltage drop across switch S₁ is reduced to zero before the turn-on driver signal is given, and zero-voltage switching for switch S₁ is achieved.

Mode E (t₄ − t₅): As depicted in Figure 3e, switches S₁ and S₂ are turned on, while switches S₃ and S₄ are turned off. As both main switches S₁ and S₂ are turned on, the input voltage is applied to the inductors L₁ and L₂, the currents i_L1 and i_L2 increase linearly, and the voltage across the primary winding of the transformer is almost zero. Then, i_L1 and i_L2 can be given as

$$i_{L1}(t)=i_{L1}(t_4)+{\displaystyle \frac{v_{\inf }-v_{Cc}}{L_1}}(t-t_4)$$

(21)

$$i_{L2}(t)=i_{L2}(t_4)+{\displaystyle \frac{v_{\inf }}{L_2}}(t-t_4)$$

(22)

3. Characteristic and Parameter Design Analysis

3.1. Current Conversion Ratio

According to the above operation mode analysis, the relationship between the average voltage values of C_r1 and C_r2 on the resonant capacitor and the average output voltage value can be given as

$$V_{cr1}=V_{cr2}={\displaystyle \frac{V_o}{2}}$$

(23)

According to (6), (7), and (23), the average voltage of the clamp capacitor can be expressed as

$$V_{Cc}={\displaystyle \frac{n{V}_o}{2}}$$

(24)

According to (24), it can be seen that the voltage of the clamp capacitor is only related to the turn ratio of the transformer and output voltage, and is independent of other parameters such as the input current or output current, so the voltage of the clamp capacitor does not need special control. As long as the output voltage is stable, the V_Cc voltage remains stable.

Based on the above operation mode analysis, for an ideal converter, the input power should be equal to the output power:

$$I_{in}{V}_{Cc}{\displaystyle \frac{L_1}{L_1+L_2}}(1-D)T_s+I_{in}{V}_{Cc}{\displaystyle \frac{L_2}{L_1+L_2}}(1-D)T_s=V_o{I}_o{T}_s$$

(25)

where D is the duty cycle of main power switch S₁ and S₂, and I_in is the input current.

According to (24) and (25), the relationship of the input current I_in and output current I_o can be written as

$$I_o={\displaystyle \frac{n{I}_{in}(1-D)}{2}}$$

(26)

3.2. Transformer Turn Ratio Design

Observing Equation (26), the design of the turn ratio of the transformer should be related to the duty cycle range, input current I_in, and output current I_o. Since the duty cycle D is usually bigger than 0.5, the minimum turn ratio condition can be written as

$$n\ge {\displaystyle \frac{2I_o}{I_{in}(1-D)}}$$

(27)

3.3. Inductance L₁ and L₂ Design

According to Kirchhoff’s law, the relationship between the currents i_L1 and i_L2 through inductors L₁ and L₂ and the input current I_in can be expressed as

$$i_{L1}+i_{L2}=I_{in}$$

(28)

The current ripple Δi_L1 and Δi_L2 of inductors L₁ and L₂ can be defined as

$$\Delta i_{L1}=i_{L1\_\max }-i_{L1\_\min }$$

(29)

$$\Delta i_{L2}=i_{L2\_\max }-i_{L2\_\min }$$

(30)

where i_{L1_max} is the maximum current on input inductor L₁, i_{L1_min} is the minimum value of the current on input inductor L₁, i_{L2_max} is the maximum current on input inductor L₂, and i_{L2_min} is the minimum value of the current on input inductor L₂.

According to the operation theory of the proposed converter, the inductor current ripples Δi_L1 and Δi_L2 meet the formula

$$\Delta i_{L1}=\Delta i_{L2}$$

(31)

According to Equations (1), (2), (18), (19) and (24), in Modes A, B, and C, the voltages v_L1 and v_L2 across inductors L₁ and L₂ can be obtained as

$$v_{L1}={\displaystyle \frac{n{V}_o{L}_1}{2\left(L_1+L_2\right)}}$$

(32)

$$v_{L2}={\displaystyle \frac{n{V}_o{L}_2}{2\left(L_1+L_2\right)}}$$

(33)

From Equations (28)–(33), neglecting the dead-time period, the inductor current ripples Δi_L1 and Δi_L2 can be obtained as

$$\Delta i_{L1}=\Delta i_{L2}={\displaystyle \frac{n{V}_o\left(1-D\right)}{2\left(L_1+L_2\right)f_s}}$$

(34)

where f_s is the switching frequency of the main power switch.

Usually, the inductors L₁ and L₂ operate in continuous conduction mode, and Δi_L1 and Δi_L2 should be smaller than the input current. Combing Equations (26) and (34), the inductance values L₁ and L₂ need to meet the following requirements:

$$L_1+L_2\ge {\displaystyle \frac{V_o{I}_o}{I_{in}{f}_s}}$$

(35)

From Equation (35), it can be observed that the sum of inductances L₁ and L₂ depends on the output voltage V_o, output current I_o, input current I_in, and switching frequency f_s. Due to the symmetrical current waveforms of inductances L₁ and L₂, they play the same role in the positive and negative half cycles. Therefore, on the basis of satisfying Equation (35), L₁ and L₂ can use the same value inductor.

3.4. Design of Clamp Capacitor C_C, Resonant Capacitor C_r, Resonant Inductor L_lk, and Magnetic Inductor L_m

In general, the value of the clamp capacitor can affect the voltage ripple ΔV_Cc of the clamp capacitor. When all output load power comes from the clamp capacitor, the voltage ripple ΔV_Cc of the clamp capacitor C_C is at maximum. Therefore, the voltage ripple ΔV_Cc should meet following equation:

$$\Delta V_{Cc}\le {\displaystyle \frac{T_s{I}_o}{2n{C}_C}}$$

(36)

To ensure the correct operation of the secondary resonant circuit, the resonant capacitor C_r and inductor L_lk should be designed to satisfy

$$\Delta v_{cr}\le V_o$$

(37)

According to the Equation (10), which is the voltage ripple expression of the resonant capacitor in the operation mode analysis of Mode B, the resonant capacitor should meet

$$C_r={\displaystyle \frac{T_s{I}_o}{2\Delta v_{cr}}}\ge {\displaystyle \frac{T_s{I}_o}{2V_{o\_\min }}}$$

(38)

where V_{o_min} is the minimum output voltage.

To guarantee zero-current switching turn-off for the rectifier diodes D₃ and D₄, the half-period π/ω_r of the resonant circuit should be less than (1 − D)T_s. The resonant inductance, which is also the secondary side leakage inductance of the transformer T, should meet following equation:

$${\displaystyle \frac{\pi }{{\omega }_r}}=\pi \sqrt{2L_{lk}{C}_r}\le \left(1-D\right)T_s$$

(39)

According to (26), Equation (39) can be rewritten as

$$L_{lk}\le {\displaystyle \frac{8}{C_r}}{\left({\displaystyle \frac{I_o{T}_s}{\pi n{I}_{in}}}\right)}^2$$

(40)

According to the operation principle and key waveform of the primary current, magnetic current, and secondary side current in Figure 4, the magnetic inductance should meet the below requirements:

$$L_m\ge {\displaystyle \frac{{\left(V_{Cc}(1-D)\right)}^2{T}_s}{2V_O{I}_O}}$$

(41)

4. Experimental Results

To verify the correctness of the proposed converter and the above theoretical analysis, a 100 W experimental prototype is built. The specifications and main circuit parameters of the proposed converter are shown in

Table 1. Key Circuit Parameters.

Design Parameter	Value
Input current Iin/A	2
Switching frequency fs/kHz	50
Output current value Io/A	1
Rated output voltage Vo/V	100
Inductance L1 and L2/mH	2
Transformer T turn ratio	2.2
Magnetic inductance Lm/μH	300
Secondary side leakage inductance Llk/μH	1.5
Resonant capacitor Cr1 and Cr2/μF	1
Clamp capacitor CC/μF	1
Main switches S1, S2, S3, S4	IRFR220NTRPBF
Schottky diode D1 and D2	MBR10200CS
Output filter capacitor Co/μF	100

. The experimental prototype and the experimental platform are presented as in Figure 5 and Figure 6, respectively. The detailed information of the experimental equipment is shown in

Table 2. Experimental equipment setup table.

Equipment Variety	Equipment Model	Function
DC Power Source 1	Chroma 62012P-600-8	Provide input power for the proposed converter.
DC Power Source 2	UNI-TREND UTP1310-II	Provide power supply for the control board.
DC Electronic load	Chroma 6314A + 63115A + 63110A	One channel provides output load for output current source, another channel works with the DC power source 1 to provide the input current source.
Oscilloscope	Tektronix DPO3014	Measure and display waveforms.
Voltage Probe	RIGOL RP1025D	Measure voltage waveforms.
Current Probe	Tektronix TCP0150	Measure current waveforms.
Digital Multimeter	Fluke 17B	Measure input voltage, output voltage, input current, and output current of experimental board.

.

Figure 7 shows the test waveforms of input current I_in, input voltage V_in, output current I_o, and output voltage V_o of the proposed converter with 100% load, 80% load, 50% load, and 10% load. From the test waveforms in Figure 8, it can be seen that the converter can convert a 2 A input current to a constant 1 A output current, even if the output voltage varies from 10 V to 100 V.

Figure 8 illustrates test waveforms of the primary voltage v_TP of the transformer, primary current i_P of the transformer, secondary current i_S of the transformer, and clamp capacitor voltage v_Cc with 100% load and 10% load. From the test waveforms, it can be seen that the clamp capacitor voltages v_Cc with 100% load and 10% load are around 110 V and 14 V, respectively, which is the same as the calculation analysis of Equation (24).

Figure 9 shows the experimental waveforms of inductor currents i_L1 and i_L2 and driver signals of S₁ and S₂ with 100% load and 10% load. It can be seen that the average current value of i_L1 and i_L2 is around 1 A, and the inductor current ripples Δi_L1 and Δi_L2 are same because of the same inductor L₁ and L₂ values. The inductor current ripples Δi_L1 and Δi_L2 with 100% load and 10% load are around 300 mA and 40 mA, respectively, which is the same as in the theoretical analysis of Equation (34). From the test waveforms of Figure 7, Figure 8 and Figure 9, the operation waveforms are the same as the theoretical analysis waveforms in Figure 4.

Figure 10 shows the experimental results of the efficiency with different output voltage loads. It can be seen that the conversion efficiency reaches 91.6% with a rated 100 V output voltage load, and the efficiency decreases to 61.5% with a 10 V output voltage load. Compared to the CS/CS converter of references [5,16], the CS/CS converters of [5,16] have no output current regulation control circuit, and thus the constant output current regulation accuracy of the proposed circuit of this paper is better than the circuit in references [5,16]. The deviation of the output current of [5,16] is greater than 8% and 4% when the output voltage changes from 100 V to 10 V, but the output current can be regulated well within the whole output voltage range. The efficiency of the circuit of reference [5] changes from 81% to 52% with the output load voltage variation from 100 V to 20 V; the efficiency of the proposed circuit of this paper changes from 91.6% to 61.5% with the output load voltage variation from 100 V to 10 V. Therefore, compared to the circuits in references [5,16], the output current regulation accuracy and conversion efficiency of the proposed circuit is improved.

5. Conclusions

By employing input current sharing and output voltage doubling circuits, an active clamp dual-inductor isolated CS/CS converter which uses a single-stage conversion circuit to realize constant current source conversion with a wide output voltage range is proposed in this paper. The active clamp circuit at the primary side of the proposed circuit can recover energy stored in leakage inductance, suppress voltage spikes of the primary side switches, and achieve zero-voltage switching of the primary side switches. The secondary side’s output voltage doubling circuit helps to achieve a wide range of output volage conversion, and the secondary side’s resonant circuit can achieve zero-current switching of the secondary side diodes, which can reduce losses and enhance efficiency. The operating principles and the characteristic and parameter design analysis, including current conversion ratio, transformer turn ratio, power inductors, and resonant capacitors and inductor, are presented. To verify the feasibility of the proposed converter, a 100 W experimental prototype was built and tested. Test results based on the experimental prototype indicated that the proposed converter can achieve current source to current source conversion with a wide output voltage variation rage, and the efficiency can reach 91.6% with a rated load.