1. Introduction
A system that can transmit radio frequency (RF) signals (
XTX(
t)) through optical fiber used to support wireless communication services, known as radio-over-fiber (RoF), has been developed recently. The transmission is completed by modulating the optical source (
Ein(
t)) using
XTX(
t), which is transmitted using the electro-optic (E/O) converter located in the central office (CO). At the receiver, the RF signal is recovered using an optoelectric (O/E) converter located in the radio access point (RAP). Recovered RF signals (
Xrec(
t)) are then transmitted wirelessly from RAP to the mobile station (MS). The conversion of RF signals to optics can be performed by modulating the optical source directly or modulating the optical carrier externally. Either direct or external modulation can be used to modulate the intensity (amplitude) or phase of the optical carrier. The recovery of the RF signal at the receiver on intensity modulation (IM) is recognized as direct detection (DD), whereas in phase modulation, it is known as coherent detection [
1].
The IM on the RoF link produces an optical signal with a double sideband spectrum. The optical double sideband (ODSB) is an optical signal with a spectrum consisting of an optical carrier and upper and lower sidebands located around the optical carrier. When the ODSB signal is transmitted through a fiber link, the chromatic dispersion of the fiber causes the sideband and optical carrier to propagate at different speeds. This leads to the modulated signal at the receiver (
ERX(
t)) experiencing a different phase between the sideband component and optical carrier by
ϕ. The proportion of
ϕ follows the length of the fiber (
L), the frequency of the RF signal (
fm) and the wavelength (
λc) used. The phase difference between the sideband and optical carrier causes the O/E process to generate two identical RF signals but with a different phase of 2
ϕ, resulting in constructive and deconstructive interference on the recovered RF signal. Destructive interference causes a reduction in recovered RF signal power, which is known as dispersion power fading (DPF). If
ϕ = π/2, the sideband cancelation effect occurs, which causes a massive loss of power or deep fade [
2].
There are several methods for overcoming the DPF, such as carrier phase shift (CPS) [
3,
4,
5,
6,
7,
8]. The CPS method uses a carrier phase shift of the
ETX(
t) signal at the transmitter. The phase of the carrier (
ϕ) is adjusted such that
ϕ at the receiver is zero. However, this method has disadvantages because
ϕ is different due to the change in
L,
fm, and
λc. Thus, it is always necessary to reset
ϕ every system change, and it is not a convenient or robust solution. Furthermore, optical carrier suppression (OCS) was proposed by [
9,
10,
11,
12,
13,
14,
15,
16]. The OCS modulation system is modulated with upper and lower sidebands but without carrier components. At the OCS,
Xrec(
t) is generated from the beating upper sideband and lower sideband with a distance of 2
fm, so the frequency of
Xrec(
t) is 2
fm. However, this method has drawbacks, such as requiring additional devices, such as downconverters, to convert
Xrec(
t) frequency to its initial value. Machine learning can also be used to overcome DPF. This method has been successfully implemented for short-range transmission [
17,
18]. However, it also has some disadvantages, one of which is a complicated configuration on the receiver side.
The reduction in DPF can be overcome using optical single-sideband (OSSB) modulation. OSSB modulation has advantages such as not requiring frequency translation [
19] or inconvenient phase receiver adjustment. OSSB modulation scheme can be generated by filtering one of the sideband modulated optical signals [
20]. This method, however, highly dependents on the optical wavelength. The OSSB modulation scheme can be produced with DD-MZM by driving one of its arms using the RF signal that is given a bias voltage ½ of the switching voltage and driving another arm using a signal with a regular phase (
θ) of 90° [
15,
21,
22,
23,
24]. DD-MZM is an E/O converter frequently used in RoF links due to its ability to generate various optical schemes with just a simple configuration.
The mathematical model of OSSB modulation is usually developed based on the Bessel function [
15]. Based on this function, the low DPF at OSSB can be obtained if the system generates the carrier signal and +1st order sideband only. To produce this condition, the OSSB modulation must be operated at low
m ≤ 0.1 because higher
m will generate more sideband orders. A higher number of sideband orders will automatically increase the DPF. However, the OSSB is only effective at low
m ≤ 0.1. However, to increase the efficiency of optic–electric conversions at the OSSB, a high
m value is needed [
22]. Therefore, it will be interesting to reduce the DPF value at high
m = 1.
In wireless communication, increasing the direct signal power [
25] and using irregular phases can reduce the effect of fading. Then, the concept of suppressing fading in wireless communication can be applied to RoF communication to suppress the effect of DPF. As a novelty, this paper proposes an asymmetric carrier divider as a direct signal with an irregular RF phase on a DD-MZ modulator for eliminating DPF, as shown in
Figure 1. In detail, the optical source generated using a laser diode (LD),
Ein(
t) is divided using an optical splitter (OS) with asymmetrical values such as power modulation (P
DD-MZM) and carrier arm (CA) power (P
CA). The signal at P
DD-MZM is modulated using DD-MZM, and the modulated signal (
EDD-MZM(
t)) is generated. The signal at the CA is added to
EDD-MZM(
t) using an optical combiner (OC), which produces the transmitted signal (
ETX(
t)). The
ETX(
t) is transmitted using fiber link
H(
f) and produces received signal (
ERX(
t)). The
ERX(
t) is detected and recovered using a photodetector (PD) and produces
Xrec(
t). This proposed method is called an asymmetric carrier divider system.
Furthermore, to further reduce the DPF value that is caused by multi-sideband order, this paper also proposes the irregular RF phase. Therefore, the optimum EDD-MZM(t) spectrum can be generated by adjusting the DD-MZM arm to produce a low DPF. The DPF level is determined using the deviation factor (DF). DF is the quantity of recovered RF signal power variation at the receiver that is measured over a certain range of fiber lengths. The smaller the DF is, the lower the DPF level or the smaller the effect of chromatic dispersion towards the RoF link.
Therefore, we combine the asymmetric carrier divider system with the irregular RF phase on the DD-MZ modulator. Therefore, the combination of the optimum value of the power ratio and the optimum value of the irregular phase will reduce DF at RoF communication significantly.
Furthermore, the significant contributions of this paper are as follows:
The laser diode (LD) power (PIN) carrier is divided asymmetrically as the power modulation (PDD-MZM) and carrier arm power (PCA). The power PCA is used to compensate for the power of the carrier of the optical field of the DD-MZM output, which is reduced due to increasing m.
The RF signal with an irregular phase (θ) was applied to the DD-MZ modulator. Therefore, the optimum EDD-MZM(t) spectrum can be generated by adjusting the DD-MZM arm.
The minimum DF is obtained when the PIN is separated as 75% for PCA and 25% for PDD-MZM with an irregular RF signal of θ = 48° and bias point value of γ = ¾. This proposed structure produces DF at m = 0.1 and m = 1 value are 0.008 and 0.03, or it reduced DF of 96.7% compared to OSSB.
The proposed system was successfully applied without additional power or filter, and the additional power or filter managed to increase the cost and complexity of RAP. Moreover, θ is independent of fiber length. Our proposed model is applied for single fiber, not for routing scenarios with many nodes.
The proposed mathematical model of the system is developed based on the Bessel function. This model can be used to evaluate all the parameters. Furthermore, the model is validated using simulation, and it has very good agreement, which validates the proposed method.
Nevertheless, the current paper has some limitations: (1) We used the comparison between numerical and simulation models without experiment due to the unavailability of devices, but this model was successfully verified and had a good agreement; (2) Our proposed method was focused on optical signals modulated by RF sine waves [
22,
26,
27,
28,
29]; (3) To eliminate the dispersion, we focused on the received RF power; (4) We assumed that the optic arm length is less than 1 m; therefore, the phase did not change significantly; (5) In this study, we investigated up to 16 scenarios, and the DD-MZM modulator was used with the best performance in the C4 scenario; and (6) As a consideration, the paper’s proposed method focused not only on asymmetric power divider but also on the optimal phase. Therefore, the minimum DF was obtained by combining the irregular phase and the power divider to reduce DF by 96.7%. We think that the global optimum DF results will also be similar to the proposed case in this scenario. Moreover, we also must consider the availability of the optical divider in the market.
3. C/N Penalty of RoF Link
The DPF of the RoF link can be seen through the carrier-to-noise (C/N) penalty curve. The C/N penalty in this paper is defined by the ratio between RF power received at distance
L or
Prec(
L) compared to the output power at MZM modulator, and it has distance
L = 0 or
. The difference value between Prec (
L) and Prec (
L = 0) occurs because of the relative phase difference between sideband and carrier. Therefore, the noise term is the phase noise. Moreover, the C/N penalty was also introduced by [
32,
33,
34]. To calculate the C/N penalty of the RoF link, it is necessary to initially model and derive the
Prec(
L) value. The RoF link model that uses DD-MZM with CA as the E/O converter can be seen in
Figure 1. The RoF link consists of DD-MZM with CA as the E/O converter, a dispersive optical fiber with
H(
f) response and photodetector (PD) as the O/E converter.
To simplify the analysis, Equation (19) can be rewritten as
where
The transfer function of the dispersive fiber link is given by [
35]
where
D is the chromatic dispersion in ps/(nm.km). The
is the optical wavelength,
f is the frequency offset of the optical carrier,
c is the speed of light in a vacuum, and
L is the fiber length. When
ETX(
t) is transmitted through the fiber link, there is a phase difference between the optical carrier and the first-order sideband,
given by
while the phase difference between the optical carrier and any random sideband is the square of the frequency range
given by [
36]
Therefore, the optical field arriving at the receiver (
ERX(
t)) can be expressed as
At the receiver,
ERX(
t) is detected using a photodetector, which is a squared envelope operator, given by [
2]
By simply taking the
fm term,
is obtained, where
Al0 =
Au0 =
Ac.
The photodetector output (
Xrec(
t)) electrical current is equivalent to the real part of Equation (26) [
35]; therefore,
The
Xrec(
t) power is the square of the amplitude term in Equation (27) [
2], so that the recovered RF signal power as the
L function,
Prec(L), is given by
where
Equation (28) shows that the recovered RF signal is obtained by multiplying the
nth order lower sideband with the (
n + 1)th order conjugated lower sideband plus the multiplication of the
nth order conjugated upper sideband with the (
n + 1)th order upper sideband, where
n is an integer. It can be seen from Equation (28) that
Prec(
L) is dependent on
γ and
θ. The C/N penalty is obtained by comparing
Prec(
L) with
Prec(
L = 0) and is given by [
2]
The DPF level in this paper is measured by using the DF given by [
32]:
where
is the i-sample of the C/N penalty,
is the average sample of the C/N penalty, and N is the number of samples. The lower the DF is, the smaller the effect of dispersion on
Prec(
L) or the better the system performance.
3.1. C/N Penalty of RoF Link with ODSB Modulation
The calculation of the C/N penalty aims to observe the DPF on the RoF link that uses DD-MZM as the E/O converter with ODSB modulation. This is done because IM on the RoF link generally produces modulated optical signals with a double sideband spectrum. Based on Equation (14), DD-MZM will generate a modulated optical signal with a double sideband spectrum only if it is set with parameters γ = ½ and θ = 180°. The following steps are completed to calculate the C/N penalty of the RoF link with the modulated ODSB. This study focused on the effect of dispersion so that attenuation was eliminated. In addition, the attenuation at optical fiber is minimal.
- (a)
Calculate Prec(L) using Equation (28) with r = 1 (without CA), Pin = 1 mw (0 dBm), γ = ½, θ = 180°, m = 0.5, λc = 1550 nm (D = 17 ps/(nm.km)), and fm = 60 GHz with L = 0, 0.1, 0.2, … 5 km.
- (b)
From the obtained Prec(L) in (a), calculate the C/N penalty using Equation (29).
- (c)
Repeat steps (a) and (b) for m = 1, 1.5, 2, 2.5, 3, 3.5, and 4.
- (d)
The result of the calculation is shown in
Figure 3a,b.
The vertical axis in
Figure 3 shows the C/N penalty in dB, whereas the horizontal axis depicts the fiber length in km. The C/N penalty of the RoF link curve with ODSB modulation at
m ≤ 2 is shown in
Figure 3a, while
Figure 3b shows the C/N penalty at
m > 2. Observed from the physical meaning side, the C/N value was influenced by constructive/destructive sideband interference. Therefore, the received power followed the cosine function, and it did not strictly increase/decrease. The RoF link with ODSB modulation at
m = 0.5, 1, and 1.5 experiences deep fade (a massive power decline) at
L = 1 and 3.1 km; at
m = 2, deep fade occurs at
L = 0.2 and 3.9 km. By using Equation (30), the DF value of the RoF link is obtained with ODSB modulation at
m = 0.5, 1, 1.5 and 2 as much as 6.5, 5.5, 4.2 and 10.5, respectively. The RoF link at
m = 2.5, 3, 3.5 and 4 undergoes deep fade at different
L and has an irregular pattern. The DFs from the RoF link with ODSB modulation at
m = 2.5, 3, 3.5 and 4 are 6.9, 5.5, 10.9, and 7.4, respectively. This shows that the level of the chromatic dispersion effect or the DPF value on the RoF link with ODSB modulation is very high.
It should be noted that several values of C/N have high positive values. This does not mean that the power of the signal was amplified. The C/N value is the ratio of the power of Prec(L) with Prec(L = 0). The C/N is positive when Prec(L) is higher than Prec(L = 0). This condition can be found when Prec(L = 0) has low power due to destructive interference and Prec(L) has high power due to constructive interference. Therefore, the C/N can be a positive or negative value.
This condition does not only apply to the irregular phase method but also to the ODSB (conventional) method. The value of
θ will affect the spectrum shape of the optic signal. Moreover, the index modulation (
m) will affect the value of
n, and the value of
n will affect the number of side bands. For instance, if we apply this equation to the ODSB (θ = 180°) system with parameter i.e., r = 1,
Pin = 1 mw (0 dBm),
γ = ½,
θ = 180°,
λc = 1550 nm (
D = 17 ps/(nm.km)), and
fm = 60 GHz, as shown in
Table 1,the value of
is different for different values of
m, and it was close to zero (very small). By this calculation, we use the different values of
. Therefore, the role of
θ is clear for the irregular phase as well as for ODSB.
3.2. C/N Penalty of RoF Link with OSSB Modulation
One of the methods used to overcome DPF is OSSB modulation. Based on Equation (14), DD-MZM can generate OSSB modulation by setting the parameters
γ = ½ and
θ = 90°. The effectiveness of OSSB modulation in overcoming DPF can be seen through the C/N penalty curve of the link. The calculation of the C/N penalty of the RoF link with OSSB modulation can be completed using steps similar to ODSB modulation but using
θ = 90°. The results of the calculation are shown in
Figure 4a,b.
Figure 4a illustrates the curve of the C/N penalty of the RoF link with OSSB modulation at
m ≤ 2.
Figure 4b shows the curve of the C/N penalty at
m > 2. There was no deep fade on the RoF link with OSSB modulation. There was almost no difference between the C/N penalty max and C/N penalty min (∆ C/N penalty) at
m = 0.1 (only 0.029 dB). The ∆ C/N penalty at
m = 0.5 is 0.560 dB, and it increased to 2.507 at
m = 1. This shows that OSSB modulation could overcome DPF effectively at
m = 0.1, and its effectiveness was reduced by increasing
m.
A comparison of the value of the ∆ C/N penalty and DF between ODSB and OSSB modulation is shown in
Table 2.
3.3. C/N Penalty of RoF Link with Irregular θ
The laser diode (LD) power distribution scheme in this paper is accomplished in 4 states, i.e.,
The whole optical signal power of the LD output is used as the DD-MZM input (basic configuration),
75% of the optical signal power generated by LD is used as the DD-MZM input,
50% of the optical signal power generated by LD is used as the DD-MZM input, and
25% of the optical signal power generated by LD is used as the DD-MZM input,
While the bias point variation (
γ) of DD-MZM is performed at 3 points, i.e.,
γ = ¼, ½, and ¾. Thus, all irregular
θ schemes analyzed in this paper can be seen in detail in
Table 3.
3.3.1. Irregular θ Values for Scheme A1, A2, A3 and A4
Irregular θ is the value chosen from 0° to 360° so that the RoF link produces the smallest DF. Irregular θ values are different for different schemes. Irregular θ values for scheme A1 are sought with the following stages:
- (a)
Calculate Prec(L) using Equation (28) with r = 1 (4:4), γ = ¼, m = 0.1, θ = 0°, Pin = 1 mw (0 dBm), λc = 1550 nm (D = 17 ps/(nm.km)), and fm = 60 GHz at L = 0, 0.1, 0.2, …, 5 km.
- (b)
Calculate the C/N penalty from the calculation results (a) using Equation (29).
- (c)
Calculate DF of all calculation results of (b) using Equation (30).
- (d)
Repeat steps (a) to (c) for θ = 1°, 2°, 3°, …, 360°.
- (e)
Find the θ value in step d), which produces the smallest DF (DFmin); this θ is called irregular θ.
- (f)
Repeat steps (a) to (e) for m = 0.2, 0.3, …, 4.
- (g)
The result of the calculation is shown in
Figure 5a–d.
The examination result curve of irregular
θ for scheme A1 is shown in
Figure 5a. The irregular
θ value on scheme A2 is searched by repeating steps (a) to (f) using
r = ¾,
r = ½ for scheme A3 and
r = ¼ for scheme A4. The examination result curve of irregular
θ for schemes A2, A3, and A4 can be seen in
Figure 5b–d.
Figure 5 reveals that for each
m used, there are two values of irregular
θ: irregular
θ I (
θI) and irregular
θ II (
θII). The different
m value and different schemes produce different irregular
θ. The minimum DF for irregular
θ I (
θI) and irregular
θ II (
θII) is shown in
Table 4.
The values of DF for each irregular
θ at 0.1 ≤
m ≤ 4 of schemes A1, A2, A3 and A4 are depicted in
Figure 6. The DF values of the RoF link with an irregular
θ at
m = 1 of schemes A1, A2, A3 and A4 are 0.089, 0.119, 0.113, and 0.070, respectively, which shows that at
m = 1, irregular
θ with scheme A4 and overcome DPF most effectively on the RoF link using DD-MZM with CA, which is biased at
γ = ¼.
3.3.2. Irregular θ Values for Scheme B1, B2, B3 and B4
An irregular
θ value for schemes B1, B2, B3 and B4 is attained by repeating the steps in finding irregular
θ values for schemes A1, A2, A3 and A4 but using
γ = ½. The curve of irregular
θ searching results for schemes B1, B2, B3 and B4, respectively, can be seen in
Figure 7a–d.
Figure 7 reveals that for each
m used, there are two values of irregular
θ: irregular
θ I (
θI) and irregular
θ II (
θII). The different
m value and different schemes produce different irregular
θ. The minimum DF for irregular
θ I (
θI) and irregular
θ II (
θII) is shown in
Table 4.
The DF values for each irregular
θ at 0.1 ≤
m ≤ 4 of schemes B1, B2, B3 and B4 are illustrated in
Figure 8. The DF values for the RoF link with an irregular
θ at
m = 1 for schemes B1, B2, B3 and B4 are 0.093, 0.524, 0.348, and 0.131, respectively, which shows that at
m = 1, irregular
θ with scheme B1 could overcome DPF most effectively on the RoF link that uses DD-MZM with CA that is biased at
γ = ½.
3.3.3. Irregular θ Values for Scheme C1, C2, C3 and C4
An irregular
θ value for schemes C1, C2, C3 and C4 is obtained by repeating the steps in finding the irregular
θ value for schemes A1, A2, A3 and A4 but using
γ = ¾. The irregular
θ search result curve for schemes C1, C2, C3 and C4 is shown in
Figure 9a–d.
Figure 9 shows that for each
m used, there are two values of irregular
θ: irregular
θ I (
θI) and irregular
θ II (
θII). The different
m value and different schemes produce different irregular
θ. The minimum DF for irregular
θ I (
θI) and irregular
θ II (
θII) is shown in
Table 4.
The DF values for every irregular
θ at 0.1 ≤
m ≤ 4 from schemes C1, C2, C3 and C4 are portrayed in
Figure 10. The DF values of the RoF link with an irregular
θ at
m = 1 for schemes C1, C2, C3 and C4 are 0.304, 1.196, 0.194, and 0.030, respectively. It clarifies that at
m = 1, irregular
θ with scheme C4 is able to overcome DPF most effectively at the RoF link that uses DD-MZM with CA biased at
γ = ¾.
3.3.4. C/N Penalty of the RoF Link with an Irregular θ for All Schemes
The performance of irregular
θ from all schemes in overcoming DPF can be observed through the DF value curve of irregular
θ in all schemes, as depicted in
Figure 11. The performance sequence of all schemes in overcoming DPF at
m = 1 from the best to the worst is scheme C4, A4, A1, B1, A3, A2, B4, C3, C1, B3, B2 and C2. DF values of schemes C4, A4, A1, B1, A3, A2, B4, C3, C1, B3, B2 and C2 is 0.030, 0.070, 0.089, 0.093, 0.113, 0.119, 0.131, 0.197, 0.309, 0.348, 0.524, and 1.196 in order. This shows that only scheme C2 has worse performance than OSSB modulation.
To observe the effectiveness of irregular
θ of all schemes to overcome DPF, the C/N penalty is calculated for all schemes at
m = 1. The calculation is performed with
Pin = 1 mw (0 dBm),
λc = 1550 nm (D = 17 ps/(nm.km)), and
fm = 60 GHz at
L = 0, 0.1, 0.2, …, 5 km. For scheme C4, r = ¼ and
γ = ¾ are used.
θI for scheme C4 at
m = 1 was 48°. For scheme A4, r = ¼ and
γ = ¼ values are employed.
θI for scheme A4 at
m = 1 is 136°. r = 1 and
γ = ¼ are used for scheme A1, and
θI for scheme A1 at
m = 1 is 141°. Furthermore, scheme B1 uses r = 1 and
γ = ½ values.
θI for scheme B1 at
m = 1 is 69°. Next, scheme A3 uses r = ½ and
γ = ¼ values.
θI of scheme A3 at
m = 1 s 137°. A2 uses r = ¾ and
γ = ¼, and it is found that
θI for the scheme at
m = 1 is 139°. Scheme B4 uses r = ¼ and
γ = ½.
θI for scheme B4 at
m = 1 is 91°. For scheme C3, the values r = ½ and
γ = ¾ are used.
θI for scheme C3 at
m = 1 is 53°. Then, the values r = 1 and
γ = ¾ are used for C1.
θI for scheme C1 at
m = 1 is 26°. For scheme B3, the values r = ½ and
γ = ½ are used.
θI for scheme B3 at
m = 1 was 90°. Scheme B2 uses the values r = ¾ and
γ = ½, and the
θI at
m = 1 is 83°. For scheme C2, the values r = ¾ and
γ = ¾ are used.
θI for scheme C2 at
m = 1 is 50°. The calculation result of this C/N penalty is illustrated in
Figure 12.
At m = 1, the power fluctuation that occurs on scheme C4 (a scheme with the smallest DF) is very small, where the difference between the maximum and minimum C/N penalties (∆ C/N penalty) is only 0.104 dB. The ∆ C/N penalty on scheme C2 (a scheme with the largest DF) reaches 3.444 dB.
4. Numerical Simulation
To evaluate the performance of irregular
θ in overcoming DPF, a comparison of the C/N penalty curve from the RoF link with ODSB modulation, OSSB, scheme C4 (best performing scheme) irregular
θ and scheme C2 (worst performing scheme) irregular
θ is carried out using OptiSystem software. An evaluation is carried out at
m = 1 with
λc = 1550 nm (D = 17 ps/nm.km) and
fm = 60 GHz. This simulation circuit is shown in
Figure 13, and the parameter setting of LiNb-MZM is shown in
Table 5.
Figure 13a is a simulation system used to measure the C/N penalty of the RoF link with ODSB and OSSB modulation. The circuit consists of a sine generator, fork 1 × 2, electrical phase shift, continuous wave (CW) laser, MZM, optical fiber, photodetector PIN, bandpass rectangle filter and electrical power meter. The sine generator functions to generate the pure RF signal. The frequency of the sine generator is set to 60 GHz. Because the switching voltage
Vπ used in the simulation is 4 V, the sine generator voltage
Vm is regulated at 1.274 V to obtain
m = 1. The sine generator output is then duplicated using fork 1 × 2. The first fork output is forwarded to the EPS and is used as an input of the MZM upper arm. The second fork output is directly used as the input of the MZM lower arm. EPS is used to shift the RF signal phase by
θ. To generate ODSB modulation, the phase shift value of EPS is regulated at 180 deg and 90 deg to produce OSSB modulation. The type MZM used is LiNb-MZM. The MZM is set with parameters as depicted in
Table 2. The MZM optical input is obtained from the CW laser. The frequency of the CW laser is set to 1550 nm, the power is set at 0 dBm, and the linewidth is set to 10 MHz. The output of the MZM is therefore transmitted through a single-mode optical fiber. The optical fiber is thus configured using the parameters shown in
Table 6.
In this simulation, the fiber attenuation effect is ignored. At the receiver, the optical signal is detected using photodetector PIN under the parameter of responsivity = 1 A/W and dark current = 10 nA. Because the output of the photodetector consists of an electrical signal with frequencies of 0, 60, 120 GHz, etc., it is filtered by means of a bandpass rectangle filter. To obtain an RF signal at 60 GHz, the parameter filter is used with a frequency of 60 GHz, bandwidth of 10 MHz, insertion loss of 0 dB and depth of 100 dB. The power of the recovered RF signal is measured using an electrical power meter. The power measurement is performed for fiber lengths of 0 to 5 km with a step of 0.1 km. The power value in the simulation is gauged in dBm. The C/N penalty value from the simulation result is obtained by subtracting Prec(L) at length L from Prec(L) at length 0.
Figure 13b is a circuit of simulations to measure the C/N penalty of the RoF link with an irregular
θ on scheme C4. The components used in this circuit are approximately similar to the simulation circuit used to measure the C/N penalty of the RoF link with ODSB and OSSB modulations. Given that the DD-MZM optical input on scheme C4 is only 25% of the LD total power, the LD power in this circuit is first divided into 4 parts using a 1 × 4 optical splitter (OS). Therefore, 1 OS output is used as the MZM optical input, and 3 OS outputs are combined using the 3 × 1 optical combiner (OC). The output of 3 × 1 OC is then recombined with the output of MZM by means of 2 × 1 OC. For scheme C4, the bias voltage 2 of DD-MZM is regulated at 3 volts, and the value of the phase shift of EPS is set at 48°. The power measurement in this simulation is identical to the measurement on links with ODSB and OSSB modulations.
Figure 13c illustrates a simulation circuit to measure the C/N penalty of the RoF link with an irregular
θ for scheme C4. The components used in this circuit are no different from the simulation circuit employed for measuring the C/N penalty of the RoF link with an irregular
θ for scheme C2. However, this simulation uses MZM optical input by 25% from the LD total power; hence, MZM is given an input of 3 × 1 OC output. An OS output is then recombined with the output of MZM using 2 × 1 OC. For scheme C2, bias voltage 2 of MZM is also set at 3 V. The comparison result between the mathematical model and simulation model has very good agreement, which shows the validity of the proposed method, as shown in
Figure 14.
The C/N penalty of the RoF is linked with ODSB modulation, and both the calculation result and simulation show a deep fade at L = 1 and 3.1 km. However, deep fade does not occur in the C/N penalty of the RoF link with OSSB modulation, although there is still a ∆ C/N penalty of 2.5 dB. The ∆ C/N penalty in the C/N penalty of the RoF link curve with an irregular θ for scheme C4 was very small.