Analysis and Implementation of a Frequency Control DC–DC Converter for Light Electric Vehicle Applications

Abstract

In order to realize emission-free solutions and clean transportation alternatives, this paper presents a new DC converter with pulse frequency control for a battery charger in electric vehicles (EVs) or light electric vehicles (LEVs). The circuit configuration includes a resonant tank on the high-voltage side and two variable winding sets on the output side to achieve wide output voltage operation for a universal LEV battery charger. The input terminal of the presented converter is a from DC microgrid with voltage levels of 380, 760, or 1500 V for house, industry plant, or DC transportation vehicle demands, respectively. To reduce voltage stresses on active devices, a cascade circuit structure with less voltage rating on power semiconductors is used on the primary side. Two resonant capacitors were selected on the resonant tank, not only to achieve the two input voltage balance problem but also to realize the resonant operation to control load voltage. By using the variable switching frequency approach to regulate load voltage, active switches are turned on with soft switching operation to improve converter efficiency. In order to achieve wide output voltage capability for universal battery charger demands such as scooters, electric motorbikes, Li-ion e-trikes, golf carts, luxury golf cars, and quad applications, two variable winding sets were selected to have a wide voltage output (50~160 V). Finally, experiments with a 1 kW rated prototype were demonstrated to validate the performance and benefits of presented converter.

1. Introduction

Clean renewable energies with power electronic techniques have been widely developed to generate alternative current (AC) voltage or direct current (DC) voltage on AC utility systems [1,2,3,4] or DC microgrid systems [5,6]. Normally, DC/DC pulse-width modulation converters (PWMs) and AC/DC PWM converters are adopted for solar power [7,8] and wind power [9] applications to convert unstable DC and AC voltage into a stable DC voltage on DC nanogrid or microgrid systems. The DC bus voltage on a DC microgrid may be 380, 760, or 1500 V for residential houses, light rail vehicles, or DC traction vehicles applications, respectively. DC–DC converters [10,11,12,13] with full bridge circuit topologies have been developed to convert low voltage (380 V) input into low voltage units (5, 12, or 48 V) for computers, server systems, light electric vehicle (LEVs), or telecommunication applications. Three level or multilevel converters [14,15,16] have been proposed for medium voltage (760 V) or high voltage (1500 V) input applications. Soft switching converters were researched in [17,18,19,20,21,22] to decrease switching losses on power semiconductors and high efficiency. Active clamp pulse-width modulation (PWM) and asymmetric PWM techniques were studied in [17,18] to improve the switching loss from half rated power to full power by adding an extra inductor on the primary side of a DC/DC PWM converter. However, the main problems of asymmetric PWM converters are unbalanced voltages and current stresses on power devices. Using the phase shift PWM technique on full bridge converters [19,20] can reduce switching loss and obtain a high circuit efficiency. The main disadvantage of phase shift PWM techniques is the hard switching problem on the lagging-leg switches. Resonant converters with the pulse frequency modulation technique were presented in [21,22] to realize low switching losses on power semiconductors over the whole load range. However, the main drawback of resonant converters is their narrow input voltage range.

To implement clean transportation alternatives and realize emission-free demands, a new PWM converter with the advantages of a wide voltage output capability and low switching loss is presented and verified in this paper for a universal battery charger in LEV or electric vehicle (EV) applications. A cascade resonant circuit was selected to lessen the voltage rating on active devices so that 600 V power switches are adopted for the 760 V input condition. Two resonant capacitors are used on the resonant tank to not only achieve resonant behavior but also input spilt voltage balance. Due to the variable switching frequency, active devices are turned on under zero voltage switching (ZVS), and the fast recovery diodes are turned off under zero current switching (ZCS). Therefore, the power losses are reduced in the presented converter. To implement the wide voltage output capability for universal battery charger applications in LEV or EV systems, two winding sets were selected for the low voltage side to have different voltage gains under high or low output voltage regions. The presented circuit has a more concise circuit configuration, less device counts, and an easier control strategy when using the general purpose PWM integrated circuit compared to the conventional DC converters. The basic DC microgrid system and the circuit configuration of the presented converter are provided in Section 2. The circuit operation for wide voltage operation capability is discussed in Section 3. The circuit analysis of the studied converter is provided in Section 4. Section 5 gives the design procedures and experiments of the presented circuit. In Section 6, the conclusion and future work directions are provided.

2. Circuit Configuration

The basic circuit blocks of DC nanogrid or microgrid systems are illustrated in Figure 1. The input voltages on a DC microgrid can be DC or AC voltage from clean renewable energies such as wind power and photovoltaic (PV) power or utility systems. Therefore, the AC/DC or DC/DC PWM converters need to change unstable AC voltage from wind turbine generators or unstable DC voltage from PV panels to stable DC voltage on DC microgrid systems. Bidirectional AC/DC or DC/DC PWM converters are needed between utility and DC microgrid systems to achieve bidirectional power flow capability. DC microgrids can supply AC motor drives, energy storage units, light rail transit systems, or battery chargers for LEVs through DC/AC inverters or DC/DC converters. The input voltage of DC transportation applications is 750 or 1500 V. However, the input voltage of a residential house or a local industry factory is 380, ±380, or 760 V. Therefore, the DC bus voltage on DC microgrids can be 380, ±380, 760, or 1500 V for different power rating requirements. High voltage PWM converters are normally adopted to supply the high power output for EV chargers or DC transportation systems.

Figure 2 provides a circuit diagram of the studied high voltage DC/DC converter. The input voltage is 760 V from the DC bus of the DC microgrid. The output voltage of the studied converter is used to charge batteries for LEV applications such as scooters, electric motorbikes, Li-ion e-trikes, golf carts, luxury golf cars, and quads. Due to the wide battery voltage range in LEV applications, DC/DC converters need to have wide output voltage operation capabilities. Figure 2a provides a diagram of the presented circuit with V_in = 760 V and V_o = 50 V–160 V. Four active devices, two input capacitors, an isolated transformer, a resonant inductor, and two resonant capacitors are used on the high voltage side (primary side). Four fast recovery diodes, an active switch, an output capacitor, and a DC resistor are adopted on the low voltage side (secondary side). Due to the series connection of four active devices on the high voltage side, active devices with a 600 V voltage rating are adopted on the primary side to withstand 760 V input. The series resonant circuit structure with L_r, C_r1, C_r2, and L_m is adopted on the presented converter to have the advantage of a soft switching operation for active devices (S₁–S₄) and fast recovery diodes (D₁–D₄). Two center-taped rectifiers with different winding turns are used on the low voltage side to extend the output voltage range operation. If switch S₅ is OFF (Figure 2b), D₁ and D₄ are OFF. The low voltage output is provided in the proposed converter with n_s secondary winding turns. The voltage gain at the low voltage output condition (Figure 2b) is V_o/V_in = G(f_sw)n_s/(4n_p), where G(f_sw) is the voltage gain of the series resonant circuit. If switch S₅ is ON (Figure 2c), D₂ and D₃ are OFF and the high voltage output is provided in the proposed converter with 2n_s secondary winding turns. Therefore, the voltage gain at the high voltage output condition is V_o/V_in = G(f_sw)n_s/(2n_p). Thus, the studied converter can provide a low (high) voltage output range with S₅ OFF (ON).

3. Circuit Operation

To realize the wide output voltage operation, the presented circuit has two equivalent sub-circuits, as shown in Figure 2b,c. S₁–S₄ are controlled with pulse switching frequency modulation (PFM). For low and high output voltage ranges, the switching signals and the main voltage and current waveforms are provided in Figure 3a,b, respectively. For the low voltage output range, the studied converter is only operated at a low voltage gain. Therefore, S₅ is off, and D₁ and D₄ are also off. In order to generate the square voltage PWM waveforms on leg voltages v_ab and v_bc, the gate waveforms of S₁ (S₂) and S₃ (S₄) are identical. If the proposed converter is used at the switching frequency f_sw < (or >) series resonant frequency f_r,1 by C_r1, C_r2, and L_r, the presented resonant converter has six (or four) working modes in every switching cycle. Figure 4 shows the equivalent working modes under f_sw < f_r,1. If the converter is operated at f_sw > f_r,1, then only modes 1, 3, 4, and 6 are used in the proposed converter for every switching period.

Mode 1 [t₀ ~ t₁]: v_CS1 = v_CS3 = 0 at time t₀. Due to i_CS3 < 0 and i_CS1 < 0, the body diodes D_S3 and D_S1 are naturally conducting. The ZVS turn-on of S₃ and S₁ is accomplished at t₀. The capacitor voltage v_Cr2 + v_Cr1 = V_dc1. Due to i_Lr (t) > i_Lm(t), the fast recovery diode D₂ is conducting and v_Lm = (n_p/n_s)V_o. L_r, C_r2, and C_r1 are resonant in mode 1 with resonant frequency $f_{r,1}=1/2\pi \sqrt{2L_r{C}_r}$ where C_r = C_r1 = C_r2.

Mode 2 [t₁ ~ t₂]: since f_r,1 < f_sw, i_D2 will decrease to 0 at t₁. Then, D₁ becomes off without reverse recovery current loss. On the primary side, L_m, L_r, C_r2, and C_r1 are naturally resonant with the resonant frequency $f_{r,2}=1/2\pi \sqrt{2(L_r+L_m)C_r}$.

Mode 3 [t₂ ~ t₃]: S₁ and S₃ are turned off at t₂, and C_S2 and C_S4 are discharged. After time t₂, i_Lm > i_Lr such that D₃ is conducting and v_Lm becomes −(n_p/n_s)V_o. The magnetizing current i_Lm(t₂) can be obtained as follows:

$$i_{Lm}(t_2)\approx (n_p{V}_o)/(4n_s{L}_m{f}_{sw})$$

(1)

The ZVS condition of S₄ and S₂ is given as:

$$i_{Lm}(t_2)\ge V_{in}\sqrt{C_{oss}/(L_m+L_r)}$$

(2)

where C_oss = C_S1 = C_S2 = C_S3 = C_S4. However, the dead time t_d between S₂ and S₁ must be greater than the discharged time of C_S2 and C_S4. Therefore, the maximum value of L_m can be obtained as follows:

$$L_m\le (n_p{V}_o{t}_d)/(4V_{in}{f}_{sw}{n}_s{C}_{oss})$$

(3)

Mode 4 [t₃ ~ t₄]: at t₃, v_CS2 = v_CS4 = 0. The body diodes D_S4 and D_S2 are naturally forward-biased. The ZVS turn-on of S₄ and S₂ is achieved. Due to i_Lr(t₃) < i_Lm(t₃), the fast recovery diode D₃ is conducting and v_Lm = −(n_p/n_s)V_o. L_r, C_r2, and C_r1 are naturally resonant on the primary side in mode 4.

Mode 5 [t₄ ~ t₅]: at time t₄, i_D3 = 0. Thus, D₃ becomes off. L_m, L_r, C_r2, and C_r1 are naturally resonant on the primary side with the resonant frequency f_r,2.

Mode 6 [t₅ ~ T_sw+t₀]: S₂ and S₄ turn off at t₅, and C_S1 and C_S3 are discharged. After time t₅, i_Lm < i_Lr such that D₂ becomes forward-biased and v_Lm = (n_p/n_s)V_o. At time t₀ + T_sw, v_CS1 = v_CS3 = 0.

For the high voltage output range, the studied converter has a high voltage gain. S₅ is turned on, and the fast recovery diodes D₂ and D₃ are reverse-biased. The presented converter has six (or four) operating modes in every switching cycle if f_sw < (or >) f_r,1. Figure 5 provides the equivalent circuits for f_sw < f_r,1.

Mode 1 [t₀ ~ t₁]: at time t₀, v_CS1 = v_CS3 = 0. D_S1 and D_S3 are conducting, v_S2,ds = v_S4,ds = V_in/2, and v_Cr1 + v_Cr2 = V_in/2. D₁ is conducting so that v_Lm = n_pV_o/(2n_s).

Mode 2 [t₁ ~ t₂]: if f_r,1 > f_sw, the i_D1 will be decreased to 0 at t₁. Then, D₁ is off.

Mode 3 [t₂ ~ t₃]: at t₂, S₁ and S₃ turn off. Due to i_Lr < i_Lm, the fast recovery diode D₄ becomes forward-biased and v_Lm = −n_pV_o/(2n_s). The magnetizing current i_Lm(t₂) can be obtained as follows:

$$i_{Lm}(t_2)\approx (V_o{n}_p)/(8f_{sw}{L}_m{n}_s)$$

(4)

The ZVS condition of S₄ and S₂ for the high voltage output region is obtained as follows:

$$i_{Lm}(t_2)\ge V_{in}\sqrt{C_{oss}/(L_m+L_r)}$$

(5)

In order to have ZVS operation, the maximum value of L_m for the high voltage output region is given as follows:

$$L_m\le (V_o{n}_p{t}_d)/(8f_{sw}{V}_{in}{C}_{oss}{n}_s)$$

(6)

Mode 4 [t₃ ~ t₄]: at t₃, v_CS2 = v_CS4 = 0. D_S4 and D_S2 become forward-biased. Due to i_Lm > i_Lr, the fast recovery diode D₄ becomes forward-biased and v_Lm = −n_pV_o/(2n_s).

Mode 5 [t₄ ~ t₅]: if f_r,1 > f_sw, the i_D4 will be decreased to 0 at time t₄. Then, D₄ becomes reverse-biased.

Mode 6 [t₅ ~ T_sw+t₀]: at time t₅, S₄ and S₂ are turned off. Due to i_Lr > i_Lm after time t₅, the fast recovery diode D₁ becomes forward-biased and v_Lm = n_pV_o/(2n_s). At time t₀ + T_sw, v_CS3 = v_CS1 = 0.

4. Circuit Analysis

For the design of the resonant converter, a general pulse frequency modulation was selected to generate the PWM signals for all power switches. Power switch S₅ is controlled by using an input voltage comparator to select low (or high) winding turns on the output side under the low (or high) voltage output region. For frequency analysis under frequency modulation, the equivalent AC circuit of the proposed converter on the primary side is given in Figure 6a. In Figure 6a, V_in,e and R_e are the equivalent AC input voltage and AC resistor on the primary side, respectively. The voltage V_in,e is a square voltage waveform with voltage values of 0 V and V_in/2. One can obtain the input voltage at fundamental frequency $V_{in,e,f}=V_{in}/(\sqrt{2}\pi )$. Components C_r1 and C_r2 are connected in parallel (C_r,eq = C_r1 + C_r2 = 2C_r) under the equivalent AC resonant tank. For the low voltage output range, the turn-ratio of the isolation transformer is n_p/n_s (S₅ off). However, the transformer turn-ratio is n_p/(2n_s) under the high voltage output range (S₅ on). One can obtain the magnetizing voltage v_Lm = ±n_pV_o/n_s (low voltage output range) or ±n_pV_o/(2n_s) (high voltage output range). The magnetizing voltage at the fundamental frequency can be obtained as $V_{Lm,f}=2\sqrt{2}{n}_p{V}_o/(n_s\pi )$ (or $\sqrt{2}{n}_p{V}_o/(n_s\pi )$) for the low output voltage range (or high output voltage range). The equivalent AC resistor R_e on the primary side is given as follows:

$$\begin{array}{l}{R}_e=\frac{8{(n_p/n_s)}^2{R}_o}{{\pi }^2}\left(\mathrm{low}\mathrm{voltage}\mathrm{output}\right)\\ \mathrm{or}\frac{2{(n_p/n_s)}^2{R}_o}{{\pi }^2}\left(\mathrm{high}\mathrm{voltage}\mathrm{output}\right)\end{array}$$

(7)

The voltage transfer function between the output and input sides is given in Equation (8) and shown in Figure 6b.

$$\begin{array}{l}\left|G\right|=1/\sqrt{{[\frac{f_n^2-1}{f_n^2}\frac{1}{l_n}+1]}^2+{(\frac{f_n^2-1}{f_n})}^2{x}^2}\\ =\frac{4n_p{V}_o}{n_s{V}_{in}}\left(\mathrm{low}\mathrm{voltage}\mathrm{output}\right)\mathrm{or}\frac{2n_p{V}_o}{n_s{V}_{in}}\left(\mathrm{high}\mathrm{voltage}\mathrm{output}\right)\end{array}$$

(8)

where $x=\sqrt{L_r/2C_r}/R_e$, l_n = L_m/L_r, and f_n = f_sw/f_r,1. From Equation (8), V_o can be obtained as follows:

$$\begin{array}{l}{V}_o=n_s{V}_{in}/[4n_p\sqrt{{[\frac{f_n^2-1}{f_n^2}\frac{1}{l_n}+1]}^2+{(\frac{f_n^2-1}{f_n})}^2{x}^2}]\left(\mathrm{low}\mathrm{voltage}\mathrm{output}\right)\\ \mathrm{or}{n}_s{V}_{in}/[2n_p\sqrt{{[\frac{f_n^2-1}{f_n^2}\frac{1}{l_n}+1]}^2+{(\frac{f_n^2-1}{f_n})}^2{x}^2}]\left(\mathrm{high}\mathrm{voltage}\mathrm{output}\right)\end{array}$$

(9)

5. Design Considerations and Experiments

In this presented converter, the following are the basic input and output electric specifications: V_in = 760 V, V_o = 50~160 V, P_o = 1000 W, f_r,1 = 100 kHz, and l_n = 7.5. For the low voltage output range, V_o = 50~90 V. For the high voltage output range, V_o is between 90 and 160 V. The circuit design procedure is based on the low voltage output range (V_o = 50~90 V). The design voltage gain of the presented converter is unity at f_sw = f_r,1 with V_in = 760 V and V_o = 50 V. From Equation (8), n_p/n_s can be obtained as follows:

$$\frac{n_p}{n_s}=\frac{V_{in}}{4V_o}=\frac{760}{4\times 50}\approx 3.8$$

(10)

The transformer T is implemented with the ferrite core (TDK EE-55) with A_e = 3.54 cm² and ΔB = 0.4 T. The assumed minimum switching frequency f_sw,min = 60 kHz at V_in = 90 V. The primary turns n_p,min can be obtained in Equation (11):

$$n_{p,\min }\ge \frac{(n_p/n_s)V_o}{2f_{sw,\min }\Delta B{A}_e}=\frac{3.8\times 90}{2\times 60000\mathrm{H}z\times 0.4T\times 354\times {10}^{-6}{m}^2}\approx 20.13$$

(11)

In this laboratory prototype, the selected winding turns were n_p = 32 and n_s = 8. Then, DC voltage gains were obtained through Equations (12) and (13):

$$G_{DC,\max ,L}=\frac{4n_p{V}_{o,\max }}{n_s{V}_{in}}=\frac{4\times 32\times 90}{8\times 760}\approx 1.89$$

(12)

$$G_{DC,\min ,L}=\frac{4n_p{V}_{o,\min }}{n_s{V}_{in}}=\frac{4\times 32\times 50}{8\times 760}\approx 1.05$$

(13)

For the high output voltage range (V_o = 90~160 V), the maximum and minimum DC gains of the presented converter could be obtained as follows:

$$G_{DC,\max ,H}=\frac{2n_p{V}_{o,\max }}{n_s{V}_{in}}=\frac{2\times 32\times 160}{8\times 760}\approx 1.68$$

(14)

$$G_{DC,\min ,H}=\frac{2n_p{V}_{o,\min }}{n_s{V}_{in}}=\frac{2\times 32\times 90}{8\times 760}\approx 0.95$$

(15)

Under the low output voltage range and the full load condition, R_e is given in Equation (16) at V_o = 90 V:

$$R_e=\frac{8{(n_p/n_s)}^2{R}_o}{{\pi }^2}=\frac{8\times {(32/8)}^2\times ({90}^2/1000)}{{3.14159}^2}\approx 105\Omega$$

(16)

The quality factor x = 0.05 and inductor ratio l_n = 7.5 were selected in this design procedure, and C_r1, C_r2, L_r, and L_m could be derived in Equations (17)–(19).

$$L_r=R_{e}x/(2\pi f_{r,1})=(0.05\times 105)/(2\pi \times 100000)\approx 8.35\mu H$$

(17)

$$C_{r1}=C_{r2}=1/(8{\pi }^2{L}_r{f}_{r,1}^2)=1/(8{\pi }^2\times 8.35\times {10}^{-6}\times {(100000)}^2)\approx 152nF$$

(18)

$$L_m=L_r\times L_n=8.35\times 7.5\approx 62.6\mu H$$

(19)

S₁–S₄ have a voltage stress of 190 V (=V_in,max/4). STF40N60M2 (650 V/22 A) power devices were selected for S₁–S₄. Die to V_o,max = 160 V, STTH15R06D (600 V/12 A) power diodes were selected for D₁–D₄. Power switch S₅ was chosen to be implemented by 6R070P6 (650 V/33 A). The selected capacitances were C_in1 = C_in2 = 330 μF/450 V and C_o = 1360 μF/200 V. The Schmitt voltage comparator with V_f = 90 V was selected to control switch S₅ (S₅ OFF if V_o ≤ 90 V or S₅ ON if V_o > 90 V). The UCC25600 was selected to implement pulse frequency modulation for S₁–S₄.

Table 1. Circuit components in the prototype circuit.

Items.	Parameter	Items	Parameter
Vin	760 V	Vo	50~160 V
Po	1 kW	S1~S4	STF40N60M2 (650 V/22 A)
Cin1, Cin2	330 µF/450 V	D1~D4	STTH15R06D (600 V/12 A)
fr,1	100 kHz	S5	6R070P6 (650 V/33 A)
Co	1360 µF/200 V	np:ns	32:8
Lr	8.35 µH	Lm	62.6 µH
Cr1, Cr2	152 nF

illustrates the circuit parameters in the laboratory prototype. Figure 7 shows a picture of the prototype circuit in the laboratory test.

The experimental results of the proposed converter for low voltage range operation are illustrated in Figure 8 and Figure 9. Figure 8 illustrates the experiments of the presented circuit under V_in = 760 V and V_o = 50 V at a 1 kW load. Figure 8a gives the experimental PWM waveforms of S₁–S₄. S₁ and S₃ were found to have the same PWM signals. In the same manner, the PWM signals of S₂ and S₄ were identical. Figure 8b shows the PWM signal of S₁ and leg voltages v_ab and v_bc. If S₁ and S₃ were on and S₂ and S₄ were off, then the leg voltages v_ab = V_Cdc1 = V_in/2 = 380 V and v_bc = 0. When S₁ and S₃ were off and S₂ and S₄ were on, the leg voltages v_ab = 0 and v_bc = V_Cdc2 = V_in/2 = 380 V. Figure 8c illustrates the experimental waveforms of v_Cr1, v_Cr2, and i_Lr. The AC voltage components of v_Cr1 and v_Cr2 were found to be complementary to each other, and the voltage value v_Cr1 + v_Cr2 = V_in/2. Due to the half bridge type resonant circuit, the DC voltage values v_Cr1,DC = v_Cr2,DC = V_in/4 = 190 V. The presented converter at V_o = 50 V had a G_DC,min,L = 1.05 theoretical DC voltage gain. The switching frequency was close to the resonant frequency. Therefore, the inductor current i_Lr was a sinusoidal waveform. Since the presented converter at V_o = 50 V was operated under the low voltage output region, S₅ was off and D₁ and D₄ were reverse-biased. Figure 8d illustrates the experimental waveforms of the secondary-side currents i_D2, i_D3, and I_o and load voltage V_o. The PWM signals of S₁ at P_o = 200 W (20% load) and 1 kW (100% load) are given in Figure 8e,f, respectively. It can be seen in Figure 8e,f that the active device S₁ turned on at ZVS from 20% load to 100% load. In the same manner, the experiments of the presented converter under V_o = 90 V and 1 kW load are illustrated in Figure 9. Following from Equations (12) and (13), the converter at V_o = 90 V output had more voltage gain than at the V_o = 50 V output condition. Thus, the switching frequency in Figure 9a–d (at V_o = 90 V output) was less than the switching frequency in Figure 8a–d (at V_o = 50 V output). For V_o = 90 V output, the theoretical DC voltage gain in Equation (12) was 1.89. Therefore, it could be expected that the switching frequency was less than the series resonant frequency shown in Figure 9c and the fast recovery diodes D₂ and D₃ could be turned off at the zero current switching shown in Figure 9d. From the experiments shown in Figure 9e,f, it is clear that S₁ turned on at ZVS from 20% load to 100% load. Similarly, the experiments of the converter under high voltage range operation are provided in Figure 10 and Figure 11. For the high voltage output operation, S₅ was on and D₂ and D₃ were reverse-biased. Figure 10 and Figure 11 illustrate the experiments for the V_o = 95 and 160 V output and 100% load conditions. Following Equations in (14) and (15), the voltage gain of converter at V_o = 95 V was close to unity and the DC voltage gain at V_o = 160 V was greater than unity. Therefore, the switching frequency of the presented converter at the V_o = 95 V (160 V) output was greater (less) than the resonant frequency. This phenomenon can be observed in Figure 10a–d and Figure 11a–d. For the V_o = 95 V condition, it can be seen that f_sw > f_r,1. In Figure 10d, the fast recovery diodes D₁ and D₄ were turned off at hard switching. On the other hand, the switching frequency f_sw < f_r,1 under the V_o = 160 V output condition. Therefore, D₁ and D₄ were turned off at zero current switching, as shown in Figure 11d. From the test experiments in Figure 10e,f and Figure 11e,f, it can be seen that S₁ was turned off at ZVS from 20% load to full load for both V_o = 95 and 160 V. Since the circuit characteristics of S₂, S₃, and S₄ are the same as S₁, it could be expected that S₁–S₄ had the soft switching turn-on operation. The experiments of output voltage V_o and switch S₅ are provided in Figure 12 during the output voltage variation between 50 and 160 V. If the output voltage was less than 90 V, S₅ was off (v_S5,g = 0 V). On the other hand, S₅ was on (v_S5,g = 15 V) if V_o > 90 V. The presented converter had 90%, 94%m and 91% efficiencies at V_o = 50, 95, and 160 V, respectively, under the V_in = 760 V and P_o = 1 kW conditions. The proposed converter at V_o = 95 V was operated at f_sw > f_r,1. Therefore, power switches were operated at ZVS turn-on and rectifier diodes were operated at hard switching turn-off. Due to the f_sw > f_r,1 at the V_o = 95 V condition, the higher switching frequency reduced the magnetizing current loss compared to the V_o = 160 V condition (f_sw < f_r,1). Therefore, the circuit efficiency at the V_o = 95 V condition was better than V_o = 160 V.

6. Conclusions

A new DC resonant converter is presented and verified in this paper in DC nano- or micro-grid systems with a wide output voltage capability. The main contributions of the presented DC–DC converter are (1) a high input voltage circuit structure that uses a flying capacitor to balance two input capacitor voltages, (2) a wide output voltage operation using a control switch, and (3) a wide ZVS operation range for wide output voltage and load ranges. Since the square voltage waveforms are generated on v_ab and v_bc, the voltage v_Cr1 + v_Cr2 = V_Cdc1 (if S₁ and S₃ are on) with d = T_sw/2 or V_Cdc2 (if S₂ and S₄ are on) with d = T_sw/2. Thus, the flying capacitors C_r1 and C_r2 can be used to balance input capacitor voltages V_Cdc1 = V_Cdc2 = V_in/2. To solve the high voltage application problems, a cascade resonant circuit is used on high voltage side to limit the voltage stress on active devices at V_in/2. Therefore, the general purpose power MOSFETs with a 600 V voltage rating can be used in the presented circuit. To realize wide output voltage requirements, such as for battery chargers for universal electric motorcycles or vehicles, the converter has two winding sets to extend the output voltage range using a control switch. In order to lessen the switching losses on active devices and increase the circuit efficiency, a series resonant converter using a LLC circuit structure was adopted for the presented converter to have a ZVS turn-on characteristic on active devices and a possible ZCS turn-off characteristic on fast recovery diodes. Therefore, the proposed converter can achieve soft switching operation from 200 to 1000 W load in the experimental test. Finally, the design procedures and experiments of the prototype circuit are provided to show the circuit performance and effectiveness of the studied converter. Further work on this circuit will consider reducing the number of circuit components and cost, as well as extending the output voltage range for universal battery charging station applications.