Frequency-Controlled Current-Fed Resonant Converter with No Input Ripple Current

Abstract

This paper studies a frequency-controlled current-fed resonant circuit. The adopted direct current (DC)-to-DC converter contains two boost circuits and a resonant circuit on the primary side. First, two boost circuits are connected in parallel to achieve voltage step-up and reduce input ripple current by using interleaved pulse-width modulation. Therefore, the size and current rating of boost inductors are decreased in the proposed converter. Second, the boost voltage is connected to the resonant circuit to realize the mechanism of the zero-voltage switching of all active switches and zero-current switching of all diodes. Two boost circuits and a resonant circuit use the same power devices in order to lessen the switch counts. The voltage doubler topology is adopted on the secondary side (high-voltage side). Therefore, the voltage rating of diodes on the high-voltage side is clamped at output voltage. The feasibility of the studied circuit is confirmed by the experimental tests with a 1 kW prototype circuit.

1. Introduction

Due to increased global warming and climbing temperature issues, renewable energy sources have been developed to produce clean energy. A fuel cell is a kind of renewable energy source that converts chemical energy to electric direct current (DC) or alternating current (AC) power. A solar cell is the other source of renewable energy to convert photovoltaic (PV) energy to electric DC or AC power. The outputs of fuel cell stacks and PV panels are low voltage. For connecting fuel cell stacks and PV panels to AC grids or DC grids [1,2], a voltage boost circuit and an isolation transformer are necessary. The high-voltage boost converters can be the voltage source type [3,4] or the current source type [5,6,7] circuit topologies. Normally, the input current ripple of the current source converters is much smaller than the voltage source converters. However, the main problems of high-voltage boost converters operating under high switching frequency are serious core losses and switching losses. Power converters with soft-switching turn-on or turn-off characteristics have been proposed and studied to overcome these problems. Duty cycle control [8,9,10,11,12,13] and frequency control schemes [14,15,16] are normally used to regulate load voltage and also reduce switching losses under zero-voltage or zero-current. The advantages of duty cycle control with fixed switching frequency are easy implementation with commercial integrated circuits and many available circuit topologies. However, the drawback of these topologies is poor circuit efficiency at low load, due to hard switching at low output power. Resonant converters [14,15,16] with frequency modulation have developed to achieve advantages of low switching losses at whole load range and high efficiency under low load conditions. Two boost converters and two full-bridge resonant converters with interleaved pulse-width modulation (PWM) are adopted in [5] to obtain high voltage gain and less input ripple current for electric vehicle (EV) and hybrid electric vehicle (HEV) applications. However, more power devices (10 power switches) are used in this circuit topology such that the cost is increased and the reliability is reduced.

This paper proposes and studies a simple voltage-boost current-fed resonant DC/DC circuit. The studied converter has two voltage boost circuits, a frequency-controlled full-bridge circuit and a voltage doubler rectifier. The voltage boost circuits and full-bridge circuit use the same active devices so that the total active devices are decreased. Interleaved PWM is adapted to control two voltage boost circuits. Due to the interleaved PWM operation with one-half cycle phase shift, the input current ripple of the proposed converter is reduced to zero. Since the full-bridge circuit is controlled by frequency modulation, the mechanism of the zero-voltage switching of all active switches and zero-current switching of all diodes can be realized at whole load range. Therefore, the turn-on switching losses of active switches and turn-off switching losses or the reverse recovery current losses of the diodes are decreased. A voltage doubler circuit topology is used on the secondary side (high-voltage side) in order to limit the voltage stress of diode at load voltage. In Section 2, the circuit diagram and structure of the studied circuit topology are presented and discussed. The circuit operation is presented in Section 3. The circuit performance and design examples are provided in Section 4. Finally, the feasibility of the developed circuit is verified by a 1 kW prototype circuit in Section 5, followed by the conclusions.

2. Proposed Converter

Figure 1a gives the circuit schematic of the studied converter. V_in is input voltage, V_o is output voltage, Q₁~Q₄ are active switches, L_b1 and L_b2 are boost inductors, C_b is boost capacitor, C_r and L_r are resonant capacitor and inductor, T is an isolation transformer, D₁ and D₂ are fast recovery diodes, and C_o1 and C_o2 are output capacitors. On the secondary side (high-voltage side), a voltage doubler rectifier is employed to limit the voltage rating of fast recovery diodes for high voltage output applications. The proposed circuit includes two interleaved boost converters (V_in, Q₁~Q₄, L_b1, L_b2 and C_b) and a full-bridge resonant converter (C_b, Q₁~Q₄, L_r, C_r, T, D₁, D₂, C_o1 and C_o2) to achieve high voltage gain, no input ripple current and soft switching of active switches. In Figure 1b, two voltage boost converters are operated by using interleaved PWM to achieve voltage step-up. Since the duty cycle of each of the active switches Q₁~Q₄ is 0.5, it can obtain V_b = 2 V_in. Since the gated signals of two voltage boost converters are phase-shifted by one-half of the switching period, the boost inductor ripple currents Δi_Lb1 and Δi_Lb1 are cancelled each other so that the input ripple current Δi_in = Δi_Lb1 + Δi_Lb2 = 0. Therefore, no ripple current (Δi_in = 0) is realized at input side. In Figure 1c, the full-bridge resonant circuit is operated by frequency control to produce a nearly sinusoidal current and voltage on the resonant tank by L_r, C_r and L_m. Due to the resonant tank of the full-bridge circuit being worked at inductive load, the zero-voltage switching of Q₁~Q₄ is realized at the whole load. Frequency modulation is used to control load voltage. The studied converter can be used in a battery charger/discharger with switches on the high-voltage terminal in the proposed converter and low-voltage input renewable energy source such as solar cell panel.

3. Circuit Operation

Two boost circuits with interleaved PWM schemes are connected in parallel to decrease the current stress of active devices and to obtain no input ripple current. Since Q₁~Q₄ all have 0.5 duty cycles, it can obtain the boost voltage V_b = 2 V_in. The full-bridge resonant circuit is operated at inductive load with frequency control. Therefore, Q₁~Q₄ have zero-voltage switching with low switching loss, and D₁ and D₂ have zero-current switching with no reverse recovery current loss. The studied converter is assumed L_b1 = L_b2 = L_b, C_Q1 = C_Q2 = C_Q3 = C_Q4 = C_Q, C_o1 = C_o2 and V_o1 = V_o2 = V_o/2. Figure 2 gives the PWM waveforms of the studied circuit at a switching cycle. Based on the switching states of D₁, D₂ and Q₁~Q₄, the studied converter can be divided into six operating steps in each switching period when the series resonant frequency is more than the switching frequency. Figure 3 gives these six equivalent circuits. Before step 1, Q₁~Q₄ are off, D₂ conducts, i_Lb1 > 0, i_Lb2 > 0 and i_Lr > 0.

Step 1 [t₀~t₁]: At time t₀, capacitors C_Q1 and C_Q4 are discharged to zero voltage. Since i_Lb1 − i_Lr < 0 and i_Lr + i_Lb2 > 0, the body diodes D_Q1 and D_Q4 are forward biased. At this time instant, switches Q₁ and Q₄ are turned on under zero-voltage switching. Since Q₁ and Q₄ are turned on in this step, v_a = 0 and v_b = V_b. The boost inductor voltages v_Lb1 = V_in and v_Lb2 = V_in − V_b. Since V_b > V_in, i_Lb1 increases linearly and i_Lb2 decreases linearly.

$$\frac{d{i}_{Lb1}}{dt}=\frac{V_{in}}{L_b}, \frac{d{i}_{Lb2}}{dt}=\frac{V_{in}-V_b}{L_b}$$

(1)

For the full-bridge resonant converter, energy is transferred from C_b to the secondary side (high-voltage side) through a resonant tank with L_r and C_r. Since D₂ is forward biased, the primary winding voltage v_Lm = −V_o/2, the current of L_m, i_Lm, decreases linearly with −nV_o/(2L_m) and C_o2 is charged from the secondary current of transformer T. In this step, L_r and C_r are resonant with $f_r=1/2\pi \sqrt{L_r{C}_r}$. The solutions of primary side current and voltage are given as

$$i_{Lr}(t)=\frac{[\frac{n{V}_o}{2}-2V_{in}-v_{Cr}(t_0)]}{\sqrt{L_r/C_r}}\sin \frac{t-t_0}{\sqrt{L_r{C}_r}}+i_{Lr}(t_0)\cos \frac{t-t_0}{\sqrt{L_r{C}_r}}$$

(2)

$$v_{Cr}(t)=i_{Lr}(t_0)\sqrt{L_r/C_r}\sin \frac{t-t_0}{\sqrt{L_r{C}_r}}+\frac{n{V}_o}{2}-2V_{in}-[\frac{n{V}_o}{2}-2V_{in}-v_{Cr}(t_0)]\cos \frac{t-t_0}{\sqrt{L_r{C}_r}}$$

(3)

where n = n_p/n_s. If the series resonant frequency f_r is more than the switching frequency f_sw, then the secondary side current of transformer T decreases to zero current at time t₁ and the circuit operation goes to step 2. Otherwise, the circuit operation will go to step 3 under f_sw > f_r condition.

Step 2 [t₁~t₂]: i_D2 decreases to zero current at time t₁ and D₂ becomes reverse biased. In this step, i_Lb1 increases linearly, i_Lb2 decreases linearly and i_Lr freewheels through Q₁ and Q₄. In this step, L_r, L_m and C_r are resonant with ${\omega }_p=1/\sqrt{(L_r+L_m)C_r}$. The solutions of primary current i_Lr and resonant capacitor voltage v_Cr in this freewheeling state are given in (4) and (5).

$$i_{Lr}(t)=-\frac{[2V_{in}+v_{Cr}(t_1)]}{\sqrt{(L_r+L_m)/C_r}}\sin \frac{t-t_1}{\sqrt{(L_r+L_m)C_r}}+i_{Lr}(t_1)\cos \frac{t-t_1}{\sqrt{(L_r+L_m)C_r}}$$

(4)

$$v_{Cr}(t)=-2V_{in}+[2V_{in}+v_{Cr}(t_1)]\cos \frac{t-t_1}{\sqrt{(L_r+L_m)C_r}}+i_{Lr}(t_1)\sqrt{(L_r+L_m)/C_r}\sin \frac{t-t_1}{\sqrt{(L_r+L_m)C_r}}$$

(5)

Step 3 [t₂~t₃]: Q₁ and Q₄ are turned off at time t₂ under zero voltage. Since i_Lb1 − i_Lr > 0 and i_Lr + i_Lb2 < 0, C_Q2 and C_Q3 are discharged linearly.

$$\frac{d{v}_{CQ2}}{dt}=-\frac{i_{Lb1}-i_{Lr}}{2C_Q}, \frac{d{v}_{CQ3}}{dt}=\frac{i_{Lr}+i_{Lb2}}{2C_Q}$$

(6)

In this step, D₁ is forward biased and the primary winding voltages v_Lm is clamped at nV_o/2. The discharge time of C_Q2 and C_Q3 is soon enough so that i_Lb1, i_Lb2 and i_Lr are almost constant during this step.

Step 4 [t₃~t₄]: The voltages on C_Q2 and C_Q3 are decreased to zero voltage at time t₃. Since i_Lb1 − i_Lr > 0 and i_Lr + i_Lb2 < 0, D_Q1 and D_Q4 are forward biased. Therefore, Q₂ and Q₃ can be turned on after t₃ and the zero-voltage switching of Q₂ and Q₃ is achieved. In this step, v_a = V_b and v_b = 0 so that v_Lb1 = V_in − V_b and v_Lb2 = V_in. Therefore, i_Lb1 decreases linearly and i_Lb2 increases linearly.

$$\frac{d{i}_{Lb1}}{dt}=\frac{V_{in}-V_b}{L_b}, \frac{d{i}_{Lb2}}{dt}=\frac{V_{in}}{L_b}$$

(7)

The energy is transferred from C_b to C_o1 through L_r and C_r. Since D₁ is forward biased, the primary winding voltage v_Lm = V_o/2, i_Lm increases linearly and C_o1 is charged. If the series resonant frequency f_r is more than the switching frequency f_sw, then the diode current i_D1 will decrease to zero at time t₄ and circuit goes to step 5. On the other hand, the circuit will go to step 6 under the f_sw > f_r condition.

Step 5 [t₄~t₅]: i_D1 decreases to zero current at t₄ and D₁ becomes reverse biased. During step 5, i_Lb1 decreases, i_Lb2 increases and i_Lr freewheels through Q₂ and Q₃. C_r, L_r and L_m are resonant in step 5.

Step 6 [t₅~t₀]: At time t₅, Q₂ and Q₃ turn off under zero voltage. Since i_Lb1 − i_Lr < 0 and i_Lr + i_Lb2 > 0, C_Q1 and C_Q4 are discharged linearly.

$$\frac{d{v}_{CQ1}}{dt}=\frac{i_{Lb1}-i_{Lr}}{2C_Q}, \frac{d{v}_{CQ4}}{dt}=-\frac{i_{Lr}+i_{Lb2}}{2C_Q}$$

(8)

Rectifier diode D₂ is forward biased and v_Lm = −nV_o/2. Since the discharge time of C_Q1 and C_Q4 in step 6 is very soon, i_Lb1, i_Lb2 and i_Lr are almost constant in this time interval.

4. Circuit Characteristics

4.1. Boost Converter

Two interleaved voltage boost circuits are worked in continuous conduction mode under the following assumptions: (1) power devices are ideal; and (2) inductors and capacitors are linear and time-invariant. Since the average value of the voltages across L_b1 and L_b2 is zero, it is possible to calculate the following equation based on the flux balance on the boost inductors.

$$V_b=V_{in}/(1-D)$$

(9)

where D is a duty ratio of Q₁ and Q₄. Since D is fixed at 0.5 in the studied circuit, it can be calculated as $V_b=2V_{in}$ and the DC voltage transfer function of boost converter is given as $M_{V. boost}=V_o/V_{in}=2$. The PWM waveforms of Q₁ and Q₃ are phase shifted by T_sw/2. The ripple currents, Δi_Lb1 and Δi_Lb1, on two boost inductors L_b1 and L_b2 are cancelled each other which gives Δi_in = Δi_Lb1 + Δi_Lb2 = 0. Thus, there is no ripple current (Δi_in = 0) from input voltage. The voltage ratings of Q₁~Q₄ are equal to V_b (=2 V_in). The average inductor currents I_Lb1 and I_Lb2 equal to P_o/(2 V_in).

4.2. Full-Bridge Resonant Converter

The full-bridge resonant circuit includes two switching legs to draw two current pulses per switching cycle from the input voltage terminal and to deliver more output power than the half-bridge resonant circuit. To obtain advantages of the good light load efficiency of the series resonant circuit and the ability to control load voltage at light or open load of the parallel resonant circuit, the full-bridge LLC resonant circuit is employed in the studied circuit topology. Fundamental harmonic analysis from Steigerwald’s article [17] is used to obtain the AC gain characteristics. Due to the PWM waveforms of Q₁~Q₄, the voltage v_ab is a square waveform with two voltage levels, V_b (=2 V_in) and −V_b (=−2 V_in). Thus, the fundamental frequency voltage v_ab,f is expressed as

$$v_{ab,f}=(8V_{in}/\pi )\sin (2\pi f_{sw}t)$$

(10)

Due the on-off states of D₁ and D₂, the transformer primary voltage v_Lm is a square waveform and the fundamental magnetizing voltage is given as

$$v_{Lm,f}=\frac{2n{V}_o}{\pi }\sin (2\pi f_{sw}t-\theta )$$

(11)

The fundamental secondary current of transformer is derived as

$$i_{s,T}=\pi I_o\sin (2\pi f_{sw}t-\theta )$$

(12)

Based on (11) and (12), R_ac is obtained from the secondary load R_o reflected into the primary side.

$$R_{ac}=\frac{v_{Lm,f}}{i_{s,T}/n}=\frac{2n^2}{{\pi }^2}{R}_o$$

(13)

Figure 4 gives the AC equivalent circuit of the adopted resonant circuit. v_ab,f is an effectively sinusoidal input voltage and R_ac is an effective load. Therefore, the gain characteristics of the studied resonant circuit under different switching frequencies is calculated as

$$G_{ac}(f_{sw})=\frac{\frac{R_{ac}\times j{\omega }_{sw}{L}_m}{R_{ac}+j{\omega }_{sw}{L}_m}}{j{\omega }_{sw}{L}_r-\frac{j}{{\omega }_{sw}{C}_r}+\frac{R_{ac}\times j{\omega }_{sw}{L}_m}{R_{ac}+j{\omega }_{sw}{L}_m}}=\frac{1}{1+K(1-\frac{1}{F^2})+jQ(F-\frac{1}{F})}$$

(14)

where f_sw is the switching frequency, $f_r=1/2\pi \sqrt{L_r{C}_r}$, $F=f_{sw}/f_r$, K=L_r/L_m and $Q=\sqrt{L_r/C_r}/R_{ac}$. The amplitude of the AC gain from (14) can be further expressed as

$$|G_{ac}(f_{sw})|=\frac{1}{\sqrt{{[1+K(1-\frac{1}{F^2})]}^2+Q^2{(F-\frac{1}{F})}^2}}$$

(15)

4.3. Design Procedure of the Developed Converter

A 1000 W prototype is set up to investigate the main circuit parameter design considerations. The electrical specifications of the developed converter are input voltage V_in = 44~52 V, output voltage V_o = 400 V, output power P_o = 1000 W, and the series resonant frequency f_r = 100 kHz. First, the boost converter with interleaved PWM is used to boost input voltage and reduce input ripple current. The duty cycle of Q₁~Q₄ is fixed at 0.5 with frequency modulation. The ripple currents Δi_Lb1 and Δi_Lb2 are calculated as

$$\Delta i_{Lo1}=\Delta i_{Lo2}=\frac{V_{in}{T}_{sw}}{2L_b}$$

(16)

Since the gate signals of Q₁ and Q₃ are phase shifted by T_sw/2 and each duty cycle of switches is equal to 0.5, it is obvious that the current ripple on the input side is decreased significantly and equals zero. The adopted boost inductors L_b1 and L_b2 are 37 µH. That means that the ripple currents on L_b1 and L_b2 at maximum input voltage and series resonant frequency are

$$\Delta i_{Lo1}=\Delta i_{Lo2}=\frac{V_{in}{T}_{sw}}{2L_b}=\frac{52\times {10}^{-5}}{2\times 37\times {10}^{-6}}\approx 7\mathrm{A}$$

(17)

In order to make sure the output voltage can be regulated at all input voltage ranges, the minimum DC voltage gain G_dc,min under V_Cb,max input is designed as unity. Thus, the turn-ratio is obtained in (18).

$$n=\frac{G_{dc,\min }{V}_{\mathrm{C}b,\max }}{V_o/2}=\frac{1\times 52\times 2}{400/2}=0.52$$

(18)

Transformer T is implemented by ferrite core TDK PC40 EER-42 with n_p = 13 and n_s = 25. Then, the theoretical maximum DC voltage gain under minimum input voltage is given as

$$G_{dc,\max }=\frac{n{V}_o}{2V_{Cb,\min }}=\frac{0.52\times 400}{2\times 44\times 2}=1.18$$

(19)

Based on (13), R_ac under full load conditions can be calculated as

$$R_{ac}=\frac{2n^2}{{\pi }^2}{R}_o=\frac{2\times {0.52}^2}{{3.1416}^2}\times \frac{{400}^2}{1000}\approx 8.77\mathrm{\Omega }$$

(20)

The inductor ratio K = L_r/L_m will affect the circulating current loss on the primary side of the resonant converter. The lower inductor ratio K can reduce the circulating current loss due to larger L_m. However, the AC gain of resonant converter is reduced. The higher inductor ratio K can obtain larger AC voltage gain. However, the circulating current loss is increased to reduce circuit efficiency. Therefore, the selection K is a compromise between the AC voltage gain and the circulating current loss. Normally, the K is selected between 0.08–0.5. Considering these factors, the inductor ratio and quality factor are designed as K = L_r/L_m = 1/6 and Q = 0.3. Therefore, L_r and C_r are calculated as

$$L_r=\frac{Q{R}_{ac}}{2\pi f_r}=\frac{0.3\times 8.77}{2\pi \times \mathrm{100,000}}\approx 4.18\mathsf{\mu }\mathrm{H}$$

(21)

$$C_r=\frac{1}{4{\pi }^2{L}_r{f}_r^2}=\frac{1}{4{\pi }^2\times 4.18\times {10}^{-6}\times {(\mathrm{100,000})}^2}\approx 606\mathrm{nF}$$

(22)

Considering the practical value of C_r, a 600 nF film capacitor is used for C_r. In a similar way, the resonant inductor L_r is actually selected as 4.22 µH. Since the inductor ratio K = L_r/L_m = 1/6 is adopted, the magnetizing inductance L_m is calculated as

$$L_m=L_r/k=\frac{4.22\mathsf{\mu }\mathrm{H}}{1/6}\approx 25.32\mathsf{\mu }\mathrm{H}$$

(23)

The voltage stress and average current of fast recovery diodes D₁ and D₂ are calculated in (24) and (25).

$$v_{D1,stress}=v_{D1,stress}\approx V_o=400\mathrm{V}$$

(24)

$$i_{D1,av}=i_{D2,av}=I_o=2.5\mathrm{A}$$

(25)

Diodes OM5262SW with 1000 V voltage rating and 12 A average current rating are adopted for D₁ and D₂. The voltage rating of Q₁~Q₄ are 2 × 52 V = 104 V. The MOSFETs IRFB52N15D with 150 V voltage rating and 60 A current rating are adopted for Q₁~Q₄. The 1100 µF capacitor is used for C_b and the 360 µF capacitors are used for C_o1 and C_o2.

5. Experimental Results

The developed converter is implemented by a 1 kW prototype to demonstrate the circuit performance and verify the feasibility and effectiveness of the studied circuit. Figure 5 gives the experimental circuit diagram of the developed circuit. The photocoupler PC817 and voltage regulator TL431 are adopted to control load voltage. The resonant mode control integrated circuit UCC25600 is used to realize frequency modulation and achieve zero-voltage switching. The circuit parameters of active and passive components of the developed circuit are discussed and obtained from the previous section. From the experimental results in Figure 6, Figure 7, Figure 8, Figure 9, Figure 10, Figure 11 and Figure 12, the test and experimental results agree with the PWM waveforms in Figure 2. The experimental results of the gate voltages of Q₁~Q₄ at rated power are illustrated in Figure 6. It can be observed that Q₁ (Q₂) and Q₃ (Q₄) have complementary PWM waveforms. The switching frequency of Q₁ at 44 V input is less than the switching frequency at 52 V input so that the resonant circuit can obtain larger voltage gain to regulate load voltage. The experimental waveforms of input current and boost inductor currents at the rated power are given in Figure 7. It can be observed that i_Lb1 and i_Lb2 are interleaved with each other and balanced well. The PWM waveforms of two boost circuits are interleaved by a phase shift of T_sw/2 so that the input ripple current Δi_in is reduced almost to zero.

The test results of boost voltage V_b, boost current i_Cb and switch currents i_Q2 and i_Q4 under the rated power are shown in Figure 8. It is clear that the frequency of i_Cb is twice the frequency of i_Q2 and i_Q4. Figure 9 gives the test results of primary voltage and current of full-bridge resonant circuit under the rated power. It can be seen that the higher switching frequency at 52 V input will result in low circulating current compared to at 44 V input.

Figure 10 illustrates the test results of i_Lb1, i_Lb2, i_Q1~i_Q4 and i_Lr under the rated power. The peak current of i_Q1 is larger than the peak current of i_Q2 due to i_Q1 = i_Lb1 − i_Lr and i_Lr < 0 when Q₁ is conducting. The test and experimental results of the output capacitor voltages and rectifier diode currents under the rated power are shown in Figure 11. V_o1 and V_o2 are balanced well and D₁ and D₂ are turned off under zero-current switching. Figure 12 illustrates the test results of switch Q₁ at 20%, 50% and 100% loads. It can be observed that zero-voltage switching of switch Q₁ is realized. Since Q₂~Q₄ have the same operation characteristics of Q₁, it can be expected that the zero-voltage switching of switches Q₂~Q₄ are also achieved. The measured efficiencies of the developed converter are 90.5% (at 20% load), 92.6% (at 50% load) and 94.5% (at 100% load) under 52 V input. The measured switching frequencies are 131 kHz (at 20% load), 118 kHz (at 50% load) and 99 kHz (at 100% load) under 52 V input.

6. Conclusions

A novel frequency-controlled current-fed resonant circuit with no input current ripple is proposed and verified in this paper. Theoretical examination and test verification demonstrate that a high-performance resonant circuit with low switching losses and input current ripple-free is achieved with the developed circuit topology. The zero-voltage switching of active devices on the low-voltage side and zero-current switching of rectifier diodes on the high-voltage side are also realized at the whole load range. The test results with the 1 kW laboratory prototype clearly provide the claimed characteristics. The constructed converter can be applied in the renewable energy conversion system with low-voltage input (48 V) and high-voltage output (400 V) with no input current ripple. The studied converter can be used in a battery charger/discharger with switches on the high-voltage terminal in the proposed converter. If the input voltage is from the PV solar cell panel, a slightly wider input voltage variation is expected. Then, the wider switching frequency range and maximum power point tracking must be implemented to regulate the load voltage.