New Miniature Narrow Band Microstrip Diplexer for Recent Wireless Communications

Abstract

Using Kappa substrate material, a compact microstrip diplexer is developed in this research with two separate channels based on the coupled junction and two bandpass filters functioning in independent frequency bands. Each filter comprises an input/output feed line and a number of resonators with different impedances. The diplexer’s frequency response was modeled and optimized using the Sonnet EM solver. At 2.84 and 4.08 GHz for TX/RX channels, the insertion loss is better than 1 dB for both channels, while the return loss values are 21.2 and 17 dB for transmit and receive filters, respectively. The microstrip diplexer has miniature dimensions of 24 mm × 18 mm with highly narrow bands and band isolation of more than 35 dB. The simulated scattering parameters are in agreement with the measured ones.

1. Introduction

Over the past few years, numerous ground-breaking advancements have been made in wireless communication systems. These include introducing ultra-wideband systems, the rapid development of wireless internet technologies like Wi-Fi and WiMAX, and mobile wireless communication systems that employ 3G and 4G innovations. More robust radio frequency components are required to keep up with the pace of development. Satellites have shifted their focus from strictly stationary communications to dynamic fields, including remote sensing, maritime applications, and mobile [1].

Various communication systems rely heavily on multiplexers. Broadband wireless communications, radio transmission, satellite communication, and so on are all examples. The most elementary multiplexer design is called a diplexer. It enables the use of a single antenna by two transmitters operating on separate channels. For various cellular radio system standards, diplexer technology is employed in both the base station and the radio handset. The diplexer features a power divider and two filters with defined frequency responses, physical dimensions, and bandwidths, as seen in Figure 1. The transmit filter must withstand the rather powerful signals generated by the transmitter. However, the receiver must be sensitive enough to notice weak signals. In addition, diplexers are commonly employed in the transceivers of broadband wireless access communication systems. Increased demand for these broadband access systems has increased the need for tiny, low-cost diplexers. These diplexers separate incoming and outgoing signals in the front end [2].

The negative group delay (NGO) has recently garnered a lot of attention, particularly in the realm of RF and microwave devices. The NGD idea is a relatively new field of electrical and electronic engineering study. It is predicated on an uncommon function: producing an output signal that appears to be generated in advance of the input signal. Due to the long delays and signal distortion that arise with positive group delay (PGD), achieving a certain quality of signal transmissions needs attention to signal phase characteristics in addition to signal amplitude characteristics. The NGD function has been used to create innovative design features of RF and microwave electronic devices, such as power dividers, filters, antennas, and amplifiers, as well as to produce experimental high-performance electronic devices [3,4].

This work shows a new small microstrip diplexer built on a Kappa substrate. This diplexer uses a variety of step impedance resonators and uniform impedance resonators along with input and output ports. As can be seen in its frequency responses, the projected diplexer exhibits effective band isolation of greater than 35 dB. It also exhibits very narrow bands at each channel, with negative group delays at several frequency values.

2. Related Work

Salah I. Yahya and Abbas Rezaei presented a new structure for a flat dual-channel microstrip diplexer. It was built at the height of 0.787 mm on a dielectric substrate and has a tiny surface area of 95.7 mm². The fractional bandwidth of the first channel is 34.3%, while the second is 38.6%. Modifying the suggested diplexer allows for the second channel to be used for broadband purposes. The value of f1 = 1.6 GHz is where the L-resonance band’s frequencies can be found. WiMAX uses a frequency of f2 = 3 GHz [5]. Patch and thin cells tailored for 0.78 GHz and 1.85 GHz for GSM requirements are the primary emphases of the structural design, as indicated in [6]. Given the symmetrical nature of the suggested resonator, it is possible to learn more about its behavior by doing investigations in both the odd and even modes. In addition to its small size, sound isolation, and return loss performance, this diplexer also offers low insertion and return loss. Compact substrate integrated waveguide (SIW) diplexers with square cavities for C-band applications are shown in a new design [7]. This diplexer functions at frequencies below the cut-off frequency of the concavity, taking advantage of the propagation of evanescent modes. The computed insertion losses for the lower and higher passbands, with their centers at 6.16 GHz and 7.72 GHz are 0.83 dB and 0.77 dB, respectively. For each high insulation passband greater than 22 dB, the suggested diplexer offers a fractional bandwidth of more than 5.1%. Due to these characteristics, the suggested diplexer is a viable option for the C-band requirement. In [8], two microstrip multiband diplexers are described. The diplexers have long feed lines and bandpass filters (BPFs) for each channel. One BPF controls all the individual channel parameters, allowing for greater freedom in the layout. Long, high-isolating microstrip feeding lines support the channel filters. A microwave diplexer was built by exchanging a dual-band BPF for two single-band BPFs [9]. In contrast to the traditional diplexer, which uses many junctions to split up the energy, this design reduces the requirement for external connectors. In [10], a high-efficiency and commercial Ku-band diplexer made up of fourth-order TX and RX Chebyshev filters is introduced along with a design and realization of a microwave diplexer using groove gap waveguide resonators. The interesting consistency between the simulation and measurement data validates the design process. In [11], the authors presented a microstrip diplexer that uses modernized H-shaped resonators to achieve small size and sound band isolation. This layout is compatible with the ISM (2.4 GHz) band and the GPS (1.575 GHz) frequencies. The diplexer was created by attaching two BPFs designed with the modified H-shaped resonator and tapped microstrip input/output lines. The paper introduces a new H-shaped BPF and provides an analytic theory for its H-shaped resonator and a model of its equivalent LC circuit. The electrical capabilities of the presented microstrip diplexer are enhanced by etching two cross-slots in the ground plane to create the passbands at low frequencies. Unfortunately, the circuit’s poor grounding means it cannot function properly on the GPS L1 or L2 frequencies. An ADS simulator has been used to analyze the diplexer’s performance, and CST microwave studio has been used to verify those results. In [12], the authors developed a microwave diplexer using substrate-integrated waveguide (SIW) technology. For simplicity and compactness, the diplexer was made using a common junction rather than the traditionally used external connection for energy distribution. The use of microstrip structures with a huge relative dielectric constant of 80 and no short-circuited elements [13] allows for the development of diplexers using two- and three-resonator microstrip structures. With frequency-contiguous channels for the differential mode and strong common-mode rejection, a new type of balanced dual-band planar diplexer was projected in [14]. It uses a diplexer’s symmetry plane, which contains filtering cells loaded with two stubs and coupled to one another through resistors. Each diplexer channel can operate as a differential-mode dual-passband filter with severe rejection at both passbands. Low common-mode in-band input power reflection and improved common-mode suppression are further benefits. Using microstrip open-loop coupled resonators, a dual-channel diplexer with two operational bands per channel was presented using a first channel of (1.424/1.732 GHz) in transmit and receive ends of (2.014/2.318 GHz) for a second channel [15]. When using load 1, the insertion loss is 1 dB, whereas when using load 2, it drops to 1.5 dB. The effective isolation of 35 dB between channels generates return loss magnitudes greater than 10 dB. The presented design is very good for wireless applications due to its straightforward topology, compact circuit size, and narrowband frequency responses.

3. Design Concept

A transmission line resonator with stepped impedances consists of two or more lines with a varied characteristic impedance that operates in TEM or quasi-TEM mode. Figure 2 depicts the two most common types of SIR, a short-circuited λg/4 resonator and an open-circuited λg/2 resonator. According to Figure 2, the quarter wavelength SIR comprises a short-ended line with characteristic impedance Z₁ and electrical length Ɵ₁ coupled to an open-ended line with characteristic impedance Z₂ and electrical length Ɵ₂. Two such elements connected through short-circuited ends, with this connection in place of grounding, might be thought of as the essential constituent of SIR and half-wavelength resonators [16,17].

The definition of this element allows us to see λg/4- and λg/2-type SIR as composed of one and two elements, respectively. The SIR is defined in terms of electrical parameters by the ratio of the impedances Z₁ and Z₂ of the transmission lines, as shown in the following equation [17]:

$$R_z=\frac{Z_2}{Z_1}$$

(1)

The input admittance of λ_g/4 SIR is equal to:

$$Y_{in}=j{Y}_2\frac{Y_2\tan {\theta }_1. \tan {\theta }_2-Y_1}{Y_2\tan {\theta }_1+Y_1\tan {\theta }_2}$$

(2)

The λg/4 resonator with a short circuit behaves like a parallel resonant circuit. For a quarter-wavelength SIR, the parallel resonance under condition Y_in = 0 is:

$$\tan {\theta }_1.\tan {\theta }_2=Y_1/Y_2=Z_2/Z_1=R_z$$

(3)

As shown in Equation (3), the resonant state of SIR can be determined by λg/2 transmission line and the impedance ratio R_z relative to typical uniform-impedance resonators (UIR). The electrical length is 2θ, and the characteristic impedance is Z₁. This is depicted for approximately UIR in Figure 3. Half-wavelength UIR is used to determine the angular resonance frequency for this resonator [17].

As a rule, dielectric substrate materials for UIR applications should have a small loss-tangent, a high permittivity, and a stable temperature. Comparing the two types of resonators again, we can conclude that the electrical length is the primary determinant of the resonance condition for UIR. SIR has additional flexibility that can be employed in new designs. Referring to Figure 4, where ${\theta }_T$ represents the resonant electrical length of the resonator, we have [17]:

$${\theta }_T={\theta }_1+{\theta }_2={\theta }_1+{\tan }^{-1}\left(\frac{R_z}{\tan {\theta }_1}\right)$$

(4)

For various values of the impedance ratio R_z, the overall electrical length for SIR is depicted based on ${\theta }_T$, as illustrated in Figure 4.

As it can be comprehended, the resonator’s overall electrical length has the highest magnitude at $R_z\ge 1$ and the lowest magnitude when $R_z\le 1$.

The condition for these highest and lowest values can be derived as:

$${\theta }_1={\theta }_2={\tan }^{-1}\sqrt{R_z}$$

(5)

The condition ${\theta }_1={\theta }_2$ stands for a specific condition that provides the highest and lowest lengths for SIR, as expressed below:

$${\theta }_{T min}={\tan }^{-1}\left(\frac{2\sqrt{R_z}}{1-R_z}\right)$$

(6)

Equation (6) has a minimum value for ${\theta }_T$ when $0<R_z<1$ and $0<{\theta }_T<\pi /2$. When comparing SIR to UIR, the critical difference is that the length of the resonators can be tuned by altering the impedance ratio R_z. This can be utilized for the same fundamental resonance frequency to design SIRs that are physically smaller than their UIR equivalents. SIRs regulate the passband of a bandpass filter’s first spurious mode, which can be employed in developing dual-band BPFs. The relationship between the frequency of the first spurious resonance and the frequency of SIR’s fundamental resonance is as follows:

$$\frac{f_s}{f_0}=\frac{\pi }{{\tan }^{-1}\sqrt{R_z}}-1$$

(7)

$$\frac{f_s}{f_0}=\frac{\pi }{2{\tan }^{-1}\sqrt{R_z}}$$

(8)

Equation (7) stands for the quarter-wavelength SIR ratio in the case of $\frac{f_s}{f_0}=3$ and $R_z=1$. Equation (8) represents a ratio of the half-wavelength SIR when $\frac{f_s}{f_0}=2$ and $R_z=1$. Both categories of resonators have their normalized spurious resonance frequencies illustrated in Figure 5.

Step impedance filters are a great choice if you are looking for an alternative to stub-based filters. They are the least performing type of filter because they have the highest and lowest practicable impedance transmission lines. They are an excellent option for tasks that do not require precise cutting. Increased isolation and a broad and deep stopband can be achieved with microstrip filters and diplexers based on varying SIRs [15].

The schematic for the designed circuit is shown in Figure 6. The entire circuit, including the substrate, takes up only 24 × 18 mm² in area, making it incredibly small and straightforward. Two BPFs operating in separate frequency ranges must be created in the design method. It should be noted that adjusting the length of the depicted UIR and SIR elements can change the filter center frequencies and bandwidths. Because of the coupled junction’s space-saving and diplexer design, it was chosen to connect the two filters. Because the required frequency response is compromised, combining the two filters in practice could be better. This necessitates thorough optimization of the entire circuit by an electromagnetic simulator.

Furthermore, separating the SIR- and UIR-based open-loop resonators from the feed lines can alter the simulated outcomes. Since the new Rogers Kappa-438 substrate (εr = 4.38, tanδ = 0.005) and the FR4 substrate (εr = 4.25, tanδ = 0.015) share the same dielectric constant (εr) and substrate height (h = 1.524 mm), they are envisioned as performance substitutes for one another. Microstrip devices’ size and scattering parameters are all affected by the loss tangent and dielectric constant, assuming a constant substrate thickness. Kappa-438 excels over FR4 with a minor loss tangent. Kappa-438 has a smaller loss tangent (0.005) than FR4 (0.015). I/O feeders are all non-symmetrical SIR varieties, but the upper and lower sections include clearly symmetrical and non-symmetrical parts of SIR varieties.

4. Results and Discussion

The described microstrip diplexer was designed, optimized, and simulated with the electromagnetic (EM) Sonnet simulator. Scattering parameters for diplexers with a Kappa-Rogers substrate, as simulated, are shown in Figure 7 and Figure 8. The transmit (Tx) filter has an insertion loss of 0.7 dB and a return loss of 21.2 dB at 2.84 GHz. The center frequency of the receive (Rx) filter is 4.08 GHz, and its insertion loss is 0.9 dB, while its return loss is roughly 17 dB. Furthermore, over a frequency range of 1 to 6 GHz, there is greater than 35 dB of band isolation between the two channels. The bandwidths ranges are 2.82–2.86 and 4.02–4.11 GHz for the Tx and Rx filters, respectively, which are highly narrow bandwidths. S11 response is a union of S22 and S33 responses as explained by Figure 8. Figure 9 depicts S11 responses for projected diplexer based on coupling space (g). As we can observe from this figure, the S11 response in the second channel becomes more effective as g decreases due to a tight coupling effect. Most changes are distinguished in the second channel, but minor S11 variations in the first channel exist, especially in the −3 dB region.

Figure 10 and Figure 11 depict the current intensity distribution for the projected microstrip diplexer. The highest values of magnetic strength are 42 and 53 A/M, which appeared in the corresponding resonators at 2.84 and 4.08 GHz for both cases, respectively. Figure 12 depicts the group delay for all S parameters for the projected microstrip diplexer. Negative group delay (GD) values are apparent in

Table 1. Negative group delay values.

S Parameter	Group Delay (ns)	Frequency (GHz)
S11	−3.824	4.08
S21	−8.765	3.5
S31	−6.688, −6.604	2.703, 2.923
S32	−2.263, 0.4766	3.469, 4.508

. The NGD does not contradict causality, which means that the diplexer may foretell the future location of the pulse based on its current position. PGD can be mitigated and its variability reduced with the use of NGD circuits.

The developed microstrip in this paper has the lowest compactness and the greatest band isolation between Tx and Rx channels as compared with [18,19,20,21,22,23,24,25], as explained by

Table 2. Comparing diplexers in this research article with the other reported studies.

Reference	Tx/Rx Center Frequencies in (GHz)	Insertion Loss (dB)	Return Loss (dB)	Fractional Bandwidths (%)	Isolation (dB)	Size (mm2)
[18]	1.7/2.5	2.35/1.96	31/45.8	6.11/7.44	$>21$	32 × 25
[19]	1.75/2.35	1.34/1.44	20/20	10/7.5	$>20$	14 × 26 (without feeders)
[20]	1.5/1.76	2.8/3.2	25/30	3.8/3.3	$>30$	37.3 × 59.1
[21] ADS Simulator	2.4/3.2	1.7/1.38	30/30	3.5/4.78	$>23$	16 × 47
[22]	1.95/2.14	1.22/1.12	15/17	3.08/2.8	37	74 × 40
[23]	2.65/5.4	1.94/2.55	$>15$	12.5/5.9	$>25$	35 × 23.16
[24]	1.85/2.5	2.05/2.15	$>15$	………	$>25$	50 × 50
[25]	2.36/5.17	2.3/2.8	$>10$	7.6/6.4	18	61.8 × 34.7
Proposed	2.84/4.08	0.7/0.9	21.2/17	1.41/2.2	$>$35	24 × 18

. Additionally, it has the lowest fractional bandwidths as compared with reported diplexers in [18,19,20,21,22,23,24,25].

Figure 13 is a photograph of the microstrip diplexer after it was fully fabricated. In order to confirm the validity of the design simulations and test the operation of the diplexer, it was run through its paces on a Vector Network Analyzer (VNA). Figure 14 displays the measured and simulated frequency responses of the proposed diplexer. The simulated and measured results are consistent with one another. There are small differences because of VNA setup factors including the size of the diplexer, the quality of the SMA soldering, and the effect of coaxial cable losses.

5. Conclusions and Future Work

In this paper, a miniaturized microstrip diplexer with dual channels is developed. This diplexer is built with a linked junction and two BPFs that operate in different frequency bands. Each filter is made with UIR, SIR, and feed lines. To determine the diplexer’s response, a Sonnet EM solver was used. The diplexer’s Tx/Rx channel resonances are 2.84 and 4.08 GHz, providing at least 35 dB of isolation between the two channels. The proposed diplexer has excellent electrical characteristics, negative group delay values, and a small size of 24 × 18 mm², making it suitable for use in various wireless systems. The bandwidths magnitudes are 40 and 90 MHz for Tx and Rx bandpass filters, respectively, which are highly narrow bandwidths. These narrow frequency responses are effective in preventing interferences in adjacent bands. This research can be extended as future work to take up one of the following suggestions. One, the diplexer presented in this study could be upgraded to triplexers in order to support three frequency bands. Two, two extra channels can be added with dual bands for each channel to get four different frequency ranges for a wireless system.