A Wideband Circularly Polarized Metasurface Antenna with High Gain Using Characteristic Mode Analysis

Abstract

This paper proposes a novel high-gain, wideband, circularly polarized (CP) metasurface (MTS) antenna. The antenna is composed of a centrally symmetric MTS and a slot-coupled feeding network. Through characteristic mode analysis (CMA), parasitic patches and mode-suppressing patches are added around the MTS to enhance the desired modes and suppress the unwanted modes. Subsequently, a feeding network that merges a ring slot with an L-shaped microstrip line is utilized to excite two orthogonal modes with a 90° phase difference, thereby achieving CP and high-gain radiation. Finally, a prototype with dimensions of 0.9λ₀ × 0.9λ₀ × 0.05λ₀ is fabricated and tested. The measured results demonstrate an impedance bandwidth (IBW) of 39.5% (4.92–7.37 GHz), a 3 dB axial ratio bandwidth (ARBW) of 33.1% (5.25–7.33 GHz), and a peak gain of 9.4 dBic at 6.9 GHz.

1. Introduction

Circularly polarized (CP) antennas, owing to their unique capabilities of generating and receiving signals at any polarization angle and effectively mitigating the effects of multipath interference, have found extensive applications in various wireless communication systems including satellite communications, global navigation satellite systems (GNSS) [1,2], radio frequency identification (RFID) [3,4], and 5G sub-6 GHz networks [5,6,7]. However, traditional CP antennas [8,9,10] frequently encounter intrinsic drawbacks, including low gain, a narrow axial ratio bandwidth (ARBW), and a high profile, which significantly constrain their performance and applicability in modern communication systems requiring high data rates and wide operational bandwidths.

In recent years, advancements in antenna technology have led to the exploration of innovative approaches to overcome these limitations. The metasurface (MTS) has emerged as a promising solution for enhancing the radiation performance of antennas, including bandwidth optimization [11,12,13], gain amplification [14,15,16], and polariza-tion conversion [17,18,19]. For instance, ref. [20] proposes a low-profile antenna with a stacked structure. The MTS comprises a 3 × 3 patch array placed above a custom-designed driving element. This configuration can excite multiple resonances in the MTS, leading to an enhanced ARBW of 10.2% and a gain of 8.1 dBic. However, there is an air gap between the MTS and the radiating patch in this design, which reduces the mechanical stability of the antenna structure. Moreover, its relatively narrow ARBW limits its application scenarios.

Additionally, slot-fed antennas have garnered attention for their compact size, low profile, and ease of integration [21,22,23], but they too face challenges in achieving wideband CP radiation with high gain. For instance, ref. [24] presents a co-polarized CP grid slot antenna. It uses inverted Z-shaped slots to excite rectangular-grid slot patches on the upper and lower planes, generating co-polarized CP with a broader ARBW of 20.1%.

Moreover, characteristic mode analysis (CMA) has emerged as a powerful tool for antenna designers, enabling a deeper understanding of the inherent electromagnetic behavior of antenna structures. By identifying and manipulating the characteristic modes of an antenna, designers can optimize its radiation characteristics, including bandwidth, gain, and polarization without requiring complex feeding networks or additional components [25,26]. In [27], by utilizing CMA, a 3 × 4 MTS structure was analyzed to identify the useful modes supported by this structure. However, it only identified the effective modes of the MTS without suppressing the interfering modes. Moreover, the feeding structure was too simple to fully unlock the potential of the antenna’s CP radiation. Consequently, it only achieved an ARBW of 19.42% and a gain ranging from 4.5 to 8 dBic within the CP operating band.

In this paper, a wideband CP antenna with high gain is introduced. Firstly, by employing CMA, two modes with orthogonal modal currents are selected as the desired modes. Then, parasitic patches and mode-suppressing patches are designed, respectively, to enhance the desired radiation modes and suppress the unwanted ones. Subsequently, a feeding network is designed to ensure efficient excitation of the selected orthogonal modes and the overall performance of the antenna. Through extensive simulations and parameter tuning, we optimize the dimensions and positions of the transmission lines and coupling slots to achieve a good impedance match and stable circular polarization over the wide frequency range. In addition, we conducted full-wave simulations using HFSS and demonstrated the effectiveness of CMA through a comparison of current density. Then, an antenna prototype is fabricated through Printed Circuit Board printing. Finally, its performance is experimentally verified through measurements, with the test results showing a good match with the simulation outcomes.

2. Structure of the Proposed Antenna

The structural diagram of the antenna is depicted in Figure 1. The overall physical size is 50 mm × 50 mm × 2.8 mm and the electrical size is 0.9λ₀ × 0.9λ₀ × 0.05λ₀. The antenna is composed of two dielectric substrates. The upper-layer dielectric substrate is Rogers4003C with a relative permittivity of 3.5 and a thickness of 2 mm. The MTS, parasitic patches, and mode-suppressing patches are printed on its upper surface, as shown in Figure 1a. The MTS consists of metal patches with dimensions of $p\times p$. Among them, eight unit patches located at the corners have a triangle with a side length of b truncated from each, and the gap between each unit patch is g. The area of the parasitic patches is half that of the metal patches, and the spacing between them and the nearest metal patch is g. The mode-suppressing patches are arranged in a triangular pattern around the four corners of the dielectric substrate. Each patch has a side length of $p_1\times p_1$, and the spacing between the patches is g. The lower-layer dielectric substrate is FR4 with a relative permittivity of 4.4 and a thickness of 0.8 mm. Its upper surface is a metal ground plane etched with a ring slot. The inner radius of the ring slot is $r_1$, and the outer radius is $r_2$, as shown in Figure 1b. The lower surface is an L-shaped microstrip feed line with a quarter-wavelength impedance transformer. The length and width of the impedance transformer are $l_2\times w_2$, and the width of the L-shaped microstrip feed line is $w_1$, as shown in Figure 1c. The two substrates of the antenna are closely attached to each other, and its side view is shown in Figure 1d. To facilitate the determination of the positional relationship between the feed line and the circular slot, its overlapping view is shown in Figure 1e.

After completing the structural design of the antenna, ANSYS HFSS 2022R2 software is employed to conduct optimization scans on the dimensional parameters with the aim of achieving the best ARBW and gain performance while maintaining a low-profile characteristic. Through a series of optimization operations, the final optimized parameters have been compiled and are presented in

Table 1. Optimized antenna geometric dimension parameters.

Parameters	Value (mm)	Parameters	Value (mm)
$W$	50	$h_2$	0.8
$g$	0.55	$l_1$	12.9
$p$	7.3	$l_2$	6.8
$p_1$	3.5	$l_3$	6
$c$	3	$l_4$	9.3
$r_1$	3.3	$w_1$	1.5
$r_2$	5.3	$w_2$	0.5
$h_1$	2

.

3. Characteristic Mode Analysis of the MTS

Recently, CMA has been utilized as a novel and effective approach for analyzing the internal working mechanisms and for designing the MTS antennas [25,26,27]. Characteristic modes are defined as a set of surface currents and radiation fields that depend only on the shape of the conducting object and are independent of any external sources. In antenna design, CMA provides an important theoretical basis and design guidance for optimizing antenna performance and facilitating a deeper understanding of antenna radiation mechanisms. The eigenvalue equations are defined as follows [28]:

$$X{J}_n={\mathsf{\lambda }}_{n}R{J}_n$$

(1)

where X and R are the imaginary and real parts of the impedance matrix, respectively, which is used to solve the electric field integral equation. ${\mathsf{\lambda }}_n$ is the eigenvalue, and $J_n$ is the eigenvector, which represents the mode current of the n-th (n = 0, 1, 2, 3,…) characteristic mode.

The characteristic angle represents the phase disparity between the tangential components of the mode current and the mode electric field, and it is mathematically defined as follows [28]:

$${\alpha }_n=180°-{\tan }^{-1}\left({\lambda }_n\right)$$

(2)

Modal significance (MS) is only related to the conductor itself, and its characteristic curve reflects the resonance potential at different frequencies. It is represented as follows [28]:

$$\mathrm{MS}={\displaystyle \frac{1}{1+{\lambda }_n}}$$

(3)

Generally, when MS > 0.707, this mode is in the resonance frequency band. To achieve CP radiation, at least two orthogonal modes need to be excited simultaneously, with these two modes having equal (or similar) MS values, a 90° difference in characteristic angle, and similar radiation patterns.

To understand the working mechanism of CP, the radiation patterns and mode distributions of the antenna’s MTS are studied using the CST 2022 simulation software. The structure of the MTS in the electromagnetic simulation software CST is shown in Figure 2. The region surrounding the antenna is set as a radiation boundary condition, and the dielectric layer of the antenna is set as a loss-free substrate material.

As illustrated in Figure 3, the mode currents for the first four modes are depicted under the conditions of a frequency of 6.9 GHz and an initial phase of 0°. The red arrows in Figure 3 specifically mark the regions of peak current density, the blue arrows denote the areas with minimum current density, and the black arrows are employed to indicate the direction of current for each mode. Additionally, the scale of current amplitude, which provides a quantitative reference for the magnitude of the currents, is conveniently placed on the right side of Figure 3.

Owing to the symmetry of the MTS, ${\mathbf{J}}_1$ (abbreviation for the current of Mode 1) which is polarized in the Y-direction and ${\mathbf{J}}_2$ which is polarized in the -X-direction are mutually orthogonal, thereby constituting a pair of orthogonal modes. Given that all currents within these modes are in phase, both modes generate broadside radiation patterns, as illustrated in Figure 4a,b. In contrast, the current distributions of ${\mathbf{J}}_3$ and ${\mathbf{J}}_4$ exhibit central symmetry. Due to the out-of-phase currents, radiation nulls emerge in the +Z direction, as demonstrated in Figure 4c,d. The consistency between the modal current distributions and the modal radiation patterns indicates that Mode 1 and Mode 2 are ideal directional radiation modes, capable of supporting the formation of high gain. Exciting either of these modes will result in a linearly polarized antenna. Given the inherent orthogonality between the two modes, simultaneously exciting them with a 90° phase difference will lead to the formation of a CP antenna. Nevertheless, Mode 3 and Mode 4 are considered undesirable modes in this context.

The modal significance values of the first four characteristic modes are shown in Figure 5. It can be observed that Mode 1 and Mode 2 form a pair of degenerate modes with the same modal significance across the entire frequency band. A pair of degenerate modes can achieve CP radiation through a specific feeding structure. However, the frequency bands of Mode 1 and Mode 2 overlap with those of the undesirable Mode 3 and Mode 4. This overlap may impede the realization of CP radiation. To address this problem, we have made two improvements to the MTS: Firstly, four parasitic patches are added at the center of each edge of the MTS. The currents on these parasitic patches are in phase with those of the adjacent unit cells, effectively broadening the resonance frequency bands of Mode 1 and Mode 2, as shown in Figure 5b. Secondly, mode-suppression patches are arranged in an inverted triangular pattern around the four corners of the dielectric substrate. These mode-suppression patches are precisely positioned near the regions where the currents ${\mathbf{J}}_3$ and ${\mathbf{J}}_4$ reach their maximum amplitudes. When excited, they indirectly weaken the currents on the edge patches through coupling effects between the metallic patches, thereby achieving mode suppression. After loading these mode-suppression patches, the overlapping frequency bands between the desired modes and the undesirable modes have been significantly reduced, as depicted in Figure 5c.

4. Antenna Feed Network

The design, optimization of the antenna feed network, as well as full-wave simulations were conducted in ANSYS HFSS. A novel feeding configuration, integrating a ring-shaped slot and an L-shaped microstrip conductor is employed to excite the antenna. The specific structural layout of this feeding network is illustrated in Figure 1b,c. Notably, the L-shaped microstrip line incorporates a quarter-wavelength impedance-transforming section, which facilitates good impedance matching between the microstrip feed line and the ring-shaped slot, thereby ensuring efficient power transmission and optimal antenna performance. By observing Figure 3, it can be found that the maximum currents of ${\mathbf{J}}_1$ and ${\mathbf{J}}_2$ as well as the minimum currents of ${\mathbf{J}}_3$ and ${\mathbf{J}}_4$ are concentrated at the central region of the MTS. Consequently, the feeding network should be designed beneath the center patch of the MTS to effectively excite the desired modes while suppressing the undesirable ones.

To validate the CMA and clearly establish the connection between it and the practical design analysis of the proposed structure, we conducted simulations using HFSS to simulate the current density distribution on the MTS after excitation. At the 6.9 GHz, the current density distribution in HFSS closely matches the characteristic current distribution in CST. Specifically, at the 0° phase, the current direction was predominantly along the y-axis, as illustrated in Figure 6a, whereas at the 90° phase, it shifted to the -x direction, as shown in Figure 6b. These current orientations closely align with the predictions of the characteristic modes presented in Figure 3, indicating that the feeding network successfully introduces the 90° phase difference required for circular polarization. This outcome not only validates the correctness of the theoretical analysis but also highlights the effectiveness of our proposed feeding configuration in achieving the desired CP radiation.

5. Parametric Study

Parametric study holds a pivotal position in the process of antenna design and optimization. It not only enables us to gain an in-depth understanding of the intrinsic relationships between antenna performance and various structural parameters but also provides a scientific basis for the final antenna design, ensuring that the antenna can achieve the desired performance metrics. In this study, we utilized HFSS to conduct extensive parameter sweeping and optimization. Through meticulous adjustment and analysis of different parameters, we ultimately achieved excellent broadband performance for the antenna. This chapter will elaborate on the learning and optimization processes of several key parameters involved in the antenna design.

(1)

Patch number optimization

The number of patches is an important parameter affecting antenna performance. In this study, we conducted simulation analyses on antenna structures with 3 × 3, 4 × 4, and 5 × 5 patch numbers, respectively, and compared them with the final proposed structure, as shown in Figure 7.

The 3 × 3 unit structure clearly exhibits the narrowest IBW and ARBW, indicating that an insufficient number of patches makes it difficult for the antenna to generate additional resonances. The 4 × 4 unit structure displays three resonance points on the S11 curve, located at 4.95 GHz, 5.4 GHz, and 6.03 GHz, respectively, and two resonance points on the axial ratio curve, located at 5.1 GHz and 7.25 GHz. However, the axial ratio in the frequency band between these two resonance points is greater than 3 dB. The 5 × 5 unit structure achieves an IBW of 5–7.25 GHz and an axial ratio within approximately 3 dB in the frequency range of 5.3–7.45 GHz, demonstrating the potential for realizing broadband circular polarization. Therefore, in this paper, based on the 5 × 5 unit structure, we removed corner patches and placed parasitic patches and mode-suppression patches. The specific effects will be presented in the next section.

(2)

Influence of parasitic patches and mode-suppression patches

As previously discussed in the preceding sections, adding parasitic patches and mode-suppression patches can effectively enhance the desired modes and suppress the undesired ones. To verify the effectiveness of CMA, the same feeding network was used to excite three antennas with different MTS structures, respectively, as shown in Figure 8. The comparison results of the S11 parameters and axial ratios of these three antennas, which are simulated using HFSS, are presented in Figure 9.

The impedance bandwidth of an antenna is mainly determined by its resonant characteristics and the matching condition between the antenna and the feed source. Therefore, in Figure 9a, the impedance bandwidths obtained for these three antenna structures remain basically unchanged. As shown in Figure 9b, the axial ratio values decrease with the improvement of the antenna structures, ultimately achieving an axial ratio below 3 dB across the entire operating frequency band (5.25–7.33 GHz). Although the reduction in the AR value is relatively limited, it must be noted that ensuring that the axial ratio stays below 3 dB throughout the desired operating frequency band is an essential requirement for a satisfactory CP antenna. We carefully adjusted the number and positional distribution of the patches and combined extensive software simulations and parameter scanning analyses to bring the axial ratio of Antenna 1 close to the critical threshold of 3 dB. Even achieving a minor improvement becomes extremely challenging at this point. This is because the factors affecting the axial ratio, such as the antenna’s geometric structure, the feeding network, and the interactions between different modes are highly sensitive and interrelated. A slight change in one parameter can have a complex and unpredictable impact on the overall axial ratio performance.

(3)

Optimization of coupling slot radius r

In the process of achieving the most effective coupling, the inner radius $r_1$ and outer radius $r_2$ of the ring slot are a pair of key parameters. They can be initially calculated using the following formula [29]:

$$f\approx {\displaystyle \frac{c}{\pi \left({\mathrm{r}}_1+{\mathrm{r}}_2\right)\sqrt{{\epsilon }_{\mathrm{eff}}}}}$$

(4)

$f$ is the designed central resonance frequency, and ${\epsilon }_{\mathrm{eff}}$ is the effective relative permittivity of the dielectric substrates on both sides of the slot. It can be seen from Figure 6 that the maximum current values of the most central patch are distributed at the edges of the patch. Therefore, the inner radius of the ring slot must be smaller than the side length of the most central patch to ensure that the coupling range of the ring slot can cover the edges of the most central patch. With the inner radius of the ring slot $r_1$ determined to be 3.3 mm, a parametric study on the outer radius of the ring slot r₂ is conducted to analyze its impact on the IBW and ARBW. As shown in Figure 10, it presents the axial ratio values of the CP antenna under different $r_2$. As expected, when $r_2$ starts to increase from 4.3 mm, the ARBW of the antenna gradually widens. However, a larger $r_2$ is not always better. After $r_2$ exceeds 5.3 mm, the axial ratio of the antenna increases, and the ARBW narrows. Therefore, setting $r_2$ to 5.3 mm is the optimal choice.

6. Fabrication and Measurement

To verify whether the actual performance of the antenna matches the simulation results, an antenna sample was fabricated and tested. The MTS and the feed network are printed on two separate dielectric substrates, as illustrated in Figure 11a,b, respectively. After printing, these two substrates were precisely aligned and tightly bonded together to form the complete antenna.

The reflection coefficient of the antenna was measured using a PNA3769 vector network analyzer sourced from Puna Technology Company, which is located in Nanjing, China. The results indicate that the IBW of the antenna is 4.92–7.37 GHz (39.5%), as shown in Figure 12a. The measured results are basically in line with the simulated ones. The axial ratio test outcomes for the antenna are depicted in Figure 12b. As observed, the antenna’s ARBW spans from 5.25 GHz to 7.33 GHz, accounting for a 33.1% bandwidth. The frequency-dependent variations in both the simulated and measured AR curves align closely. Notably, the antenna demonstrates two minima in its axial ratio at 5.4 GHz and 7.1 GHz, where the axial ratios decrease to 1.6 and 1.1, respectively. This indicates that the antenna possesses good CP performance. Figure 13 presents the frequency-dependent characteristics of the antenna’s gain and radiation efficiency. Within the CP operational bandwidth, the gain of the antenna exhibits a range from 7 dBic up to 9.4 dBic. Meanwhile, its radiation efficiency shows a fluctuation range between 80% and 94%. Specifically, at the frequency of 6.9 GHz, the antenna reaches its peak gain of 9.4 dBic. The gain fluctuation arises from multiple factors. On one hand, there is an inherent trade-off between improving gain and reducing axial ratio when adjusting the physical dimensions of antenna elements. For instance, modifying the size or spacing of radiating patches to enhance gain might disrupt the delicate balance required for achieving the desired axial ratio across the CP bandwidth, leading to inconsistent gain performance. On the other hand, at higher-order resonances within the CP bandwidth, the antenna’s radiation pattern degrades, causing non-optimal radiation and thus gain variations [12].

The far-field radiation patterns of the antenna are depicted in Figure 14. Specifically, Figure 14a,b presents the radiation patterns in the E-plane and H-plane, respectively, at 5.4 GHz, while Figure 14c,d show the radiation patterns in the E-plane and H-plane, respectively, at 7.1 GHz. The measured radiation patterns are basically consistent with the simulated ones. At 5.4 GHz, the cross-polarization ratio is less than −20 dB; at 7.1 GHz, it is less than −15 dB, indicating that the antenna exhibits distinct left-hand circular polarization characteristics.

To elucidate the spatial radiation characteristics of the antenna more clearly, Figure 15a displays 3D gain distribution at 5.4 GHz. It can be observed that the antenna features a unidirectional radiation beam and low back radiation, indicating high mode excitation efficiency. Figure 15b presents the 3D radiation pattern at 7.1 GHz, demonstrating stable directional radiation characteristics within the upper hemisphere. Despite the presence of minor sidelobes, a peak gain of 9.36 dBic is achieved. A comprehensive analysis of both the 2D and 3D radiation patterns reveals that the antenna exhibits good properties, including high directivity, angular stability, and symmetric beam shape. These findings confirm that the designed feeding network can effectively excite Mode 1 and Mode 2.

Table 2. Performance contrast of the proposed antenna against prior research antennas.

Ref.	Antenna Type	$\mathbf{Size}\left({\mathsf{\lambda }}_0\right)$	IBW (%)	ARBW (%)	Efficiency (%)	Cross-Polarization (dB)	Peak Gain (dBic)
[12]	patch-fed	0.58 ×0.58 × 0.056	45.6	23.4	>90%	<−20	7.6
[13]	cavity-backed	1.75 × 1.75 × 0.25	66.3	44.7	Not mentioned	<−15	6
[20]	patch-fed	0.70 × 0.70 × 0.03	20.48	10.2	>88%	<−18	8.1
[24]	slot-fed	0.95 × 0.85 × 0.2	38.32	20.1	60–75%	<−15	3.5
[27]	CPW-slot-fed	1 × 1 × 0.058	25.08	19.42	>96%	<−20	8
Proposed	slot-fed	0.9 × 0.9 × 0.05	39.5	33.1	80–94%	<−15	9.4

offers a performance comparison between the antenna proposed in this study and other wideband antennas reported in recent years, with the comparison table encompassing crucial parameters such as antenna type, ARBW, physical dimensions, and gain. Compared to traditional MTS antenna such as [12], the proposed antenna significantly improves both the ARBW and gain. The literature [13] proposes a cavity-backed antenna in which an MTS is employed in the design as a reflector to achieve a wider bandwidth. However, this comes at the cost of a large size, high profile of 0.25 ${\mathsf{\lambda }}_0$, and low gain of 6 dBic, which restricts its application in space-constrained scenarios such as automotive telematics control unit. A low-profile MTS antenna in [20] with an etched slot achieves a gain of 8.1 dBi, but has a narrow ARBW of only 10.6%, and there is an air gap between the MTS and the radiating patch in this design, which reduces the mechanical stability of the antenna structure. Reference [24] proposes an antenna where upper and lower symmetrically slotted apertures simultaneously feed the upper and lower MTSs. It achieves an ARBW of 20.1%. However, its gain is restricted to just 3.5 dBic. Reference [27] proposes a single-layer coplanar waveguide (CPW) fed MTS antenna based on the CMA. This antenna is built upon a 3 × 4 MTS structure and a rotated CPW feeding line. Both the MTS and the feeding line are relatively simple in design. It achieves an IBW of 25% and an ARBW of 19.42%. Within the CP operating bandwidth, its gain ranges from 4.5 to 8 dBic. We also employ the CMA, but we take it a step further. On one hand, we add parasitic patches and mode-suppressing patches around the MTS. These additional elements are utilized to enhance the desired modes and suppress the unwanted ones. This method enables us to have more precise control over the antenna’s radiation characteristics. On the other hand, for the feeding mechanism, we adopt a slot-coupled feeding network that combines a ring slot and an L-shaped microstrip line. This complex feeding network is designed to achieve better mode excitation and control, which is importance for improving the antenna’s gain and bandwidth. As a result, our design achieves wider IBW, ARBW, and higher gain compared to Reference [27], while both the physical and electrical dimensions of the profile height are smaller than those in [27].

7. Conclusions

This paper presents a novel high-gain, wideband MTS antenna with CP radiation. The antenna consists of a centrally symmetric MTS structure and a slot-coupled feeding network. Through CMA, parasitic patches and mode-suppressing patches are added around the MTS to enhance the desired radiation modes and suppress the unwanted ones. Subsequently, a feeding network combining a ring slot with an L-shaped microstrip line is designed to excite two orthogonal modes with a 90° phase difference, enabling wideband CP and high-gain radiation. The experimental results demonstrate that the proposed antenna achieves an IBW of 39.5% (4.92–7.37 GHz), a 3 dB ARBW of 33.1% (5.25–7.33 GHz), and a peak gain of 9.4 dBic at 6.9 GHz. In terms of practical applications, given its low-profile, high-gain, and wide bandwidth characteristics, the proposed antenna holds significant potential. In satellite communications, the low-profile design allows for seamless integration on satellites with limited space, and the high gain and wide bandwidth ensure reliable and high-speed data transmission over long distances. For radar systems, the wideband CP radiation enhances target detection and tracking capabilities, especially in complex electromagnetic environments. Moreover, in unmanned aerial vehicles (UAVs), the compact size and high-performance radiation of the antenna make it well-suited for on-board communication and sensing tasks, thereby improving the overall functionality and performance of UAVs.