Low-Profile VHF Antenna Based on Quarter-Mode Substrate-Integrated Waveguide Structure

Featured Application

Low-profile antenna for terrestrial wireless communication.

Abstract

A traditional whip antenna is suitable for terrestrial wireless communication due to its omnidirectional radiation pattern which is capable of transmitting and receiving signals from all directions. However, the length of a whip antenna is long for the very high frequency (VHF) band applications; for example, the ideal length is 80 cm at 90 MHz, which restrains the mobility when mounted on a vehicle. Many studies have been conducted to develop a low-profile antenna that can substitute a whip antenna but most of them are for relatively high frequency (GHz) applications. In this paper, a planar low-profile antenna operating in the VHF band is proposed. The height of the antenna is only 9 cm by utilizing a quarter-mode substrate-integrated waveguide (QMSIW) structure with shorting pins at the edges. The latter promotes omnidirectional radiations despite its short height. The antenna design was optimized by 3D full-wave simulation software. Subsequently, an antenna prototype was fabricated and tested. The results demonstrated that the antenna resonated at 86 MHz with a maximum gain of 1.7 dB along the azimuth direction.

1. Introduction

A whip antenna is widely used in a wireless communication system due to its simple structure and omnidirectional radiation pattern [1]. In particular for a terrestrial communication system, using an omnidirectional radiating antenna is necessary because a receiver (or transmitter) can be located in any direction from a transmitter (or receiver) along the ground at a distance. Figure 1 shows a whip antenna mounted on a vehicle for military terrestrial communications at the high-frequency (HF) or very high frequency (VHF) band [2,3]. The whip antenna is long compared to the vehicle size, since its length is proportional to the wavelength of operation frequency. For instance, the ideal length is 375 cm at 20 MHz (HF band) and 83 cm at 90 MHz (VHF band). Such a high profile not only makes a vehicle more likely to be exposed to opponents but also degrades its mobility. In this context, a whip antenna is often pulled and bent as in Figure 1b on the move. However, this can result in a poor antenna performance due to the change in the antenna impedance and radiation pattern.

Contrary to a whip antenna, a patch antenna [4] has a planar low-profile form factor, thus is easy to install on a vehicle roof. Nevertheless, it is not preferable for terrestrial communications as its radiation pattern is directional toward the zenith. Figure 2 shows a 3D radiation pattern of a patch antenna mounted on a vehicle. The patch antenna consists of a lower ground plane and an upper radiator, and they are separated by the air. It was modelled and its radiation pattern was calculated using full-wave simulation software (Ansys^® HFSS). As can be seen in Figure 2, most of the electromagnetic (EM) energy is radiating vertical to the antenna plane.

To overcome the above-mentioned limitations of whip and patch antennas, researchers have developed several patch-like low-profile antennas exhibiting a whip-like omnidirectional radiation pattern. One solution is the so-called monopolar wire-patch antenna [5,6,7,8] which incorporates shorting pins (i.e., vias) connecting the upper radiator and the lower ground. By adjusting the patch size, the strong currents flowing along the feed and shorting pins become in-phase, thus the EM energy is radiating in all directions, presenting a whip-like omnidirectional radiation pattern. They were implemented at 5.8 GHz [5], 4 GHz [6], 2.45 GHz [7], 4.1 GHz [8], which are relatively high frequencies compared to the VHF band. Another solution is to utilize a substrate-integrated waveguide structure (SIW) [9,10,11]. SIW is an EM waveguiding structure which emulates a rectangular waveguide in a planar form, e.g., a printed circuit board (PCB). A SIW has arrays of vias forming a fictional magnetic wall. With this, the antenna’s patch size can be reduced to half. Ref. [10] reported an SIW-based antenna operating at 5.9 GHz and [11] designed an SIW wearable antenna at 2.5 GHz.

In this paper, we designed a planar patch-like antenna that can be mounted on a vehicle’s roof based on an SIW structure. Different from previous SIW-based antennas operating at GHz frequency ranges, the proposed antenna was designed for the VHF band, more specifically 88 MHz, which is approximately 100 times lower than the GHz range. This required a careful optimization of the antenna geometry other than a simple scaling. In addition, the proposed antenna adopted the monopolar wire-patch antenna design by adding shorting pins in the SIW structure to generate omnidirectional radiation patterns. Followed by the simulation studies, an antenna prototype was fabricated and tested to demonstrate its feasibility. The size of the antenna was 125 cm × 125 cm × 9 cm which is 0.37λ × 0.37λ × 0.026λ in terms of the wavelength at 88 MHz. The measured antenna’s reflection coefficient was less than −10 dB at the target frequency and the measured received power showed a similarity to the simulated gain.

2. Substrate-Integrated Waveguide (SIW)

As mentioned in the introduction, an SIW structure can be used to miniaturize a planar antenna. Figure 3 shows a section of SIW [9], which has two rows of vias at the edges connecting two parallel metal plates separated by a dielectric substrate. An SIW exhibits a propagation characteristic similar to a rectangular metallic waveguide if the vias are closely spaced, thus no leakage radiation occurs.

The design parameters of an SIW are the height of substrate (h), via’s diameter (d), spacing (s), and width ($w$) as depicted in Figure 3. There are empirical design equations for an SIW derived from the rectangular waveguide theory [11]. The SIW width $w$ to guide the fundamental transverse electric mode (TE₁₀ mode) can be calculated by

$$w=w_r-1.08\frac{d^2}{s}+0.1\frac{d^2}{w_r} ,$$

(1)

where $w$_r is the width of a rectangular waveguide retaining the equivalent propagation characteristic as the SIW. Moreover, d and s are the diameter of the via and the spacing between vias, respectively. In order to avoid radiation leakages, the following requirements for d and s should be met [12]:

$$d<\frac{{\lambda }_{\mathrm{g}}}{5} \mathrm{and} s<2d$$

(2)

where ${\lambda }_{\mathrm{g}}$ is the guided wavelength. As long as Equation (2) is met, the TE₁₀ mode will be tightly guided inside the SIW since the densely lined-up via fences cut off the leaking waves.

3. Quarter-Mode SIW Antenna Design

Having a properly designed SIW that guides the TE₁₀ mode, an SIW-based antenna can be designed and then further miniaturized by taking advantage of the symmetry of the TE₁₀ mode [10,13,14]. Figure 4a shows the E-field amplitude distribution inside a section of an SIW. As can be seen, the E-field is symmetric along the transverse and longitudinal axes. Next, Figure 4b shows the E-field distribution when the SIW section is cut in half. The open boundary condition along the transverse axis forms a pseudo-magnetic wall due to the large ratio of the SIW’s width to height. With this virtual wall, the E-field distribution hardly changes, providing a half-mode SIW (HMSIW) [15,16]. In addition, a quarter-mode SIW (QMSIW) can be realized by cutting the HMSIW along the longitudinal axis where the E-field is symmetric. This is described in Figure 4c.

Based on the QMSIW structure, the proposed antenna was designed, and its geometry is shown in Figure 5. Two copper plates (patch and ground) were separated by air. The size of the patch corresponded to the size of QMSIW at 88 MHz determined by Equation (1). There were three different kinds of cylindrical metallic posts in between the patch and ground: feeding probe, shorting pins, and via fence. The feeding probe was the inner conductor of a coaxial cable brought to the radio frequency (RF) source, while the shorting pins were directly connecting the patch and ground. The outer and inner conductors of the coaxial probe were electrically connected to the lower ground and upper patch, respectively. The diameter of the shorting pins was 1 mm, which corresponded to a gauge size of 18. Their locations were carefully chosen relative to the feeding probe location to induce in-phase currents, thus promoting an omnidirectional radiation pattern [17]. By adding the shorting pins, the antenna resonant frequency was shifted lower, thus the antenna size parameters (W, L) were adjusted to shift it back. The QMSIW structure was formed by the via fence and two magnetic walls as depicted in Figure 5b. The via’s spacing and diameter were determined by Equation (2).

Table 1. QMSIW antenna parameters.

Parameter	Value (cm)	Parameter	Value (cm)
L	125	xf	62
W	125	yf	73
h	9	xp	6.2
s	5	yp	33.3
d	2.5

lists the geometrical parameters and their values, which were finalized after optimizing the initial model using the full-wave EM simulation tool, Ansys HFSS. Several key parameters such as the ground and patch size, feeding probe and shorting pin locations, numbers of vias and its diameter were optimized one by one to reach the dimensional values in Table 1.

Figure 6 shows a parametric simulation study on the reflection coefficient (S11) by varying the height, h. The S11 was calculated in the frequency range from 60 to 120 MHz. One can observe that the S11 is getting lower at the frequency of interest, 88 MHz, as h increases from 6 cm to 9 cm, while other parameters remain the same. This implies a better impedance matching condition is achieved due to an increase of the shunt inductance and a decrease of the shunt capacitance by increasing h [18]. We set the height h = 9 cm, which is approximately 0.025 wavelength at 88 MHz to achieve a low-profile form factor.

Figure 7 shows the effect of the shorting pins on the antenna’s radiation pattern. There are 2D radiation patterns of the antenna along the azimuth plane with no, one, and two shorting pins. A typical radiation pattern of a broadside radiating patch antenna can be observed when there is no shorting pin associated. That is, most of the EM energy is radiating vertical to the antenna plane similar to Figure 2. By adding one and then two shorting pins, the radiated EM energy is diverged toward the azimuth direction, making the antenna pattern more omnidirectional and suitable for the terrestrial wireless communication.

Figure 8 summarizes the 3D and 2D radiation patterns of the optimal two-shorting-pin QMSIW antenna. The simulated 3D radiation pattern in Figure 8a shows that the EM energy is horizontally distributed around the antenna plane. The calculated maximum gain is 1.7 dBi. The 2D plots in Figure 8b are along the ϕ = 0° and ϕ = 90° planes. They both show evenly distributed EM energy along the horizontal direction.

4. Antenna Prototype Fabrication and Measurement

A prototype of QMSIW antenna was fabricated based on the optimized design. Figure 9 shows the fabricated prototype with insets showing the feed, vias, and shorting pins connecting the bottom and top copper plates. We used aluminum cylinders to form the via fence, otherwise copper pins were used for the shorting pins. Styrofoam spacers were inserted to maintain the height between the copper plates. Styrofoam is known to have similar electromagnetic material properties as air [19].

The reflection coefficient (S11) of the fabricated prototype was measured using a vector network analyzer (Anritsu MS2038C) and its result was compared to the simulated S11 as illustrated in Figure 10. The test cable of vector network analyzer was connected to the antenna feed via an N-type coaxial connector. They had a good agreement except a slight difference in the resonant frequencies and minimum S11 values, 86.1 and 88 MHz, −12.9 and −15.94 dB for the measurement and simulation, respectively.

After verifying the fabricated antenna’s resonant frequency, we proceeded to measure the radiation performance of the antenna. We noted that the measurement of the antenna gain of the fabricated prototype was difficult because of the large antenna size, which was hard to mount in an antenna measurement chamber. Alternatively, an in-house experiment setup as depicted in Figure 11 was used to measure the antenna’s received power instead of the antenna gain. A 1.5 m long whip antenna (RH-10M) was used as a transmitting (Tx) antenna. It was connected to a signal generator (R&S SGT100A) whose waveform was controlled by a notebook. The fabricated QMSIW antenna was connected to a spectrum analyzer (Anritsu MS2038C) and the receiving power was recorded by sweeping the frequency from 60 to 100 MHz and by rotating the antenna 360 degrees along with the azimuth plane.

The measured result is shown in Figure 12. The black line is the measured received power from 60 to 100 MHz with a 5 MHz interval. In this experiment, the Tx power from the signal generator was set to 0 dBm and the distance between the Tx and Rx antennas were 10 m. As can be seen, the received power increases as the frequency increases and then peaks at 86 MHz, which corresponds to the resonant frequency of the QMSIW antenna. After 86 MHz, the received power drops dramatically. We compare it with the simulated antenna gain which is the red line in Figure 12. More specifically, the red line is the simulated peak gain at θ = 90° (i.e., azimuth plane). It also increases as the frequency increases and peaks at the simulated resonant frequency, 88 MHz. However, it gradually decreases after 88 MHz, which is different from the steep decrease in the measured received power. The latter may be caused by fabrication errors making the prototype antenna’s very inductive after the resonant frequency, thus the EM energy was trapped inside the antenna cavity. The received power at 86 MHz by rotating the fabricated antenna is plotted in Figure 13 (black line). It is compared with the simulated radiation pattern at θ = 90° (red line). Both are normalized to the maximum azimuth angle ϕ = 270°. It can be observed that the measured received power pattern shrinks about 0.5 dB from the simulated antenna gain pattern, but the overall patterns are similar to each other.

5. Conclusions

A low-profile omnidirectional vehicle antenna operating in the VHF band was designed, optimized, fabricated, and measured. The proposed antenna adopted a QMSIW structure for a size miniaturization and was implemented with two shorting pins at the edges for omnidirectional radiations. The antenna was optimized using the full-wave simulation tool Ansys HFSS, and then fabricated and tested in the lab. The dimension of the antenna was 125 cm × 125 cm × 9 cm, which was 0.37λ × 0.37λ × 0.026λ in terms of the wavelength at 88 MHz. The measured results showed that the reflection coefficient was −12.9 dB at the resonant frequency 86 MHz, which was slightly shifted from the simulation result but showing an overall good agreement. The S11’s <−10 dB bandwidth was 83.3–86.1 MHz. The measured azimuth radiation pattern was also similar to the simulated one, but its level was 0.5 dB lower. However, the omnidirectional radiation could be observed in spite of the short height of 9 cm (0.026λ at 88 MHz). The proposed antenna can be a candidate to substitute a long monopole antenna at the VHF frequency range at the price of a large mounting area.