2.2.2. Ultra-High Frequency Antenna
In the power transformers, the registration of the radio signals in the VHF/UHF band generated by the partial discharges is carried out with the use of antennas mounted in special dielectric windows [
36,
37] or antennas put in the tank interior through oil valves [
38,
39]. There are also new solutions known, in which, antennas are placed outside the tank and detect the leakage of electromagnetic waves through the gap (insulation gasket between the upper or lower ladle cover and the main tank) [
40] or bushing taps [
41]. Antennas placed in the dielectric windows usually allow to obtain the highest sensitivity in PD detection. The obstacle in the global popularization of this solution is the fact that the new power transformers are still very rarely equipped with dielectric windows. The factory installation of the interior UHF antennas takes place exclusively on demand of the purchaser. Additionally, the exchange of the revision hatches in the transformer being exploited to mount dielectric windows and antennas is an expensive, time-consuming, and complicated operation, both regarding technical and logistic issues. Unfortunately, the remaining solutions, i.e., antennas placed in the tank interior through oil valves and external antennas, regarding numerous limitations are not a valuable alternative [
39]. The first ones are simple monopole antennas of weak amplification and a very narrow transmission band. Additionally, the shielding effect of the oil drain valve and unfavorable placement inside the tank (usually it is a sidewall of the tank, near the bottom), have a negative impact of their efficiency, because they may hamper the registration of the radio signals generated by PDs [
42]. In turn, the external antennas are vulnerable to the influence of external electromagnetic disturbances, mainly corona from HV transmission lines. A significant problem is also the fact that the signal registered by the antenna may have very low energy because of the EM wave attenuation on the way from the PD source to the antenna [
40]. To minimize the problem, it is necessary to apply advanced algorithms of signals denoising, filtration, and recognition. It is worth mentioning that only a small part of the useful signal gets out through gaps (if they are present in the given tank construction at all) and bushing insulators—outside the transformer tank. Having considered all advantages and disadvantages of the available methods of UHF signals registration, finally, it was decided that for the needs of the monitoring system the antenna adjusted to installation in the dielectric window would be elaborated. High EM signals detection sensitivity, which directly translates into high effectiveness of the PD monitoring system operation, determined its choice.
In the work of [
21] the complex review of antennas applied for PD detection in the VHF/UHF band was presented. Among the commercial solutions, circular plate antenna (also called UHF disk sensor) prevails, since it has a simple construction and because it may be easily adjusted to the dimensions of the dielectric window, which usually is round-shaped. A disadvantage of this construction is a narrow bandwidth and the possibility to regulate the resonant frequency practically only by changing the diameter
d of the circular plate (
d = λ/4 where λ is the wavelength). Therefore, in many research centers, the works on the elaboration of the antenna not only of compact dimensions but also of high gain and wide bandwidth, which covers the frequency band of the EM signals generated by PD in oil-paper insulation system are carried out. Among the most common wideband antenna constructions proposed by researchers, dedicated to PD detection are meander-line antenna [
43], logarithmic spiral antenna [
44], different types of Archimedean spiral antennas [
45,
46], Vivaldi antenna [
47], microstrip patch antennas [
48,
49,
50], bio-inspired antennas [
51,
52], and fractal antennas such as Hilbert curve fractal antenna [
53,
54], H-fractal antenna [
55], Peano fractal antenna [
56], or Minkowski fractal antenna [
57].
From the viewpoint of efficient PD detection in a power transformer, the key antenna parameter is properly adjusted bandwidth. The research results presented in [
58] showed that PDs in oil-immersed power transformers generate EM pulses of frequencies from 150 to 700 MHz, wherein most of their energy was transmitted in a band of lower frequencies (up to 400 MHz). In turn, in [
59] the authors registered EM signals simultaneously with the use of three ultra-wideband antennas, which jointly covered a wide measurement band from 20 MHz up to 18 GHz. The research results showed that the main PD types in paper-oil insulation (PD in oil, PD in gas bubbles in oil, inter-turn discharges, surface discharges and creeping sparks on pressboard) generate EM pulses, for which the majority of the energy is transmitted in the band from 250 to 390 MHz. For that reason, it was decided that the designed antenna would be optimized to operate in this band.
In search of the most efficient and scalable antenna design, the authors tested different constructions of wideband antennas, among others logarithmic spiral antenna, circular-cross antenna, H-fractal antenna, Hilbert curve fractal antenna, and broadband planar inverted-F antenna (PIFA) [
53,
55]. The results of tests and simulations showed that the highest performance and possibility of relatively easy bandwidth regulation is offered by the modified constructions of a conventional PIFA. It is currently the most popular antenna in the world applied in mobile devices [
60,
61,
62]. This is due to its numerous advantages such as compact size compared to monopole antenna, high gain in both vertical and horizontal states of polarization, close to omnidirectional radiation pattern, and low manufacturing cost.
Figure 4 presents a schematic diagram of the conventional PIFA, which consists of a ground plane, a rectangular radiating element of length equal to the quarter-wave, feeding pin and shorting plate (or pin) to connect the ground plane with the top patch.
Resonant frequency
fr of conventional PIFA may be approximately estimated with the use of Equation (1), in which
c is the speed of light in vacuum,
εeff is the effective permittivity of the dielectric substrate,
L1 and
L2 are the width and length of the rectangular radiating element,
H is the thickness of the substrate, and
W is the width of shorting plate, assuming that
L1 >
W > 0 [
63,
64]. In order to more accurately estimate the resonant frequency of PIFA, some authors propose more complex formulas [
65].
The major limitation of standard PIFA is the narrow bandwidth. Simple methods for widening the bandwidth of PIFA are based on reducing the size of the ground plane and the use of a substrate with the low value of dielectric constant [
64]. Another technique is the employment of specially designed slits in front patch [
66,
67,
68,
69,
70], ground plane [
71], or in both these structures [
72,
73,
74]. A more advanced method of bandwidth widening is based on the use of stacked patches, often also equipped in slits [
75,
76,
77,
78,
79]. Other tuning techniques are based on varying the position of feeding pin, changing the capacitance value of the shorting plate, or changing the widths of feed and shorting plates [
80,
81]. By applying a properly selected chip resistor or chip capacitor load in place of the shorting plate one may reduce the resonant frequency of the antenna [
82].
For the needs of the monitoring system, it was decided to apply PIFA construction with the meandered radiating structure loaded with chip resistor [
83]. Meandering the radiating patch allows to reduce the dimensions of the conventional PIFA and obtain large gain [
84,
85,
86,
87,
88]. In turn, applying the chip resistor causes not only lowering of the resonant frequency but also widens the bandwidth (even several times compared to standard PIFA) [
83]. A schematic diagram of a meandered PIFA is presented in
Figure 5.
In the initial stage of designing, the necessity to adjust the antenna to the dielectric window dimensions was taken into account (
Figure 2) of internal diameter equal to 15 cm and the need to equip the antenna in a protective housing. Since the gain of PIFA increases as the ground plane size increases [
89], it was assumed that the largest possible diameter equaled 14 cm. The remaining dimensions of the prototype meandered PIFA (
Table 3) were optimized through iterative simulations performed in CST Microwave Studio 2016 (Dassault Systèmes SE, Vélizy-Villacoublay, France)
.The results of the simulation proved that the application of chip-resistor in place of the shorting pin allows to increase the bandwidth (
Figure 6), whereas by changing the distance between the patch and ground plane one may regulate the resonant frequency of the antenna (
Figure 7). Satisfying results were obtained for chip-resistor with resistance
R = 5.4 Ω and distance
H = 5 mm, for which the simulated voltage standing wave ratio was VSWR = 1.038 (S
11 = -36.34 dB), resonant frequency
fr = 347 MHz, and the −6 dB bandwidth was 73 MHz. The use of chip-resistor of higher resistance allows to widen the bandwidth even more (e.g., for
R = 7.2 Ω, the −6 dB bandwidth grew to 90 MHz). Unfortunately, it is at the expense of VSWR growth and a slight change of resonant frequency
fr, which is shifted to the lower frequencies. In turn, change in distance
H between the patch and ground plane influences the resonant frequency to a greater extent than the bandwidth. The results of the simulation proved that for
H = 3 mm the resonant frequency was 319 MHz, and for
H = 6 mm it increased to 357 MHz, wherein −6 dB bandwidth was, respectively, equal to 47 and 56 MHz.
Based on the simulation results, the prototype construction of meandered PIFA of dimensions presented in
Table 3 was manufactured. The antenna was equipped in cylindrical protective housing made from polytetrafluoroethylene (PTFE) and matched to the cavity of the active dielectric window. The antenna with the active dielectric window, inspection hole cover, and the module for conditioning and digital processing of PD pulses constitutes a complete PD sensor labeled as ISO-002-ADW/UHF (
Figure 8).
Having installed the antenna in the active dielectric window, the VSWR characteristic was measured using a vector network analyzer type KC901S+ (Deepace, Guangdong, China). The measurements showed that compared to the simulation results, the VSWR curve is slightly shifted towards lower frequencies (
Figure 9). It is probably an effect of the antenna surroundings influence, i.e., metal transformer tank and inspection hole cover, which were not taken into account in the simulations. The measured resonant frequency of the antenna was equal to 329 MHz (VSWR = 1.13; S
11 = −24.3 dB), and −6 dB bandwidth equal to 73 MHz (from 280 to 353 MHz).
In order to check the performance of meandered PIFA, comparative research was conducted, in which a commercial UHF disk sensor with a diameter of 15 cm and measured resonant frequency
fr = 475 MHz was used as a reference antenna. Both sensors were installed in an oil-filled transformer tank of dimensions 1200 × 800 × 750 mm, equipped in two ceramic dielectric windows and an electrode system for generating surface discharges on the pressboard sample. The distance between the PD source and the central point of the antenna, in the case of both sensors, was the same and equal to about 45 cm. The pulses were registered with a four-channel oscilloscope type Tektronix MDO 3104 with a sampling frequency of 5 GS/s. The level of the apparent charge of the generated partial discharges was monitored by the use of the standard measurement set compliant with the IEC 60270 standard [
90] (
Figure 10).
The research results confirmed the high sensitivity of the prototype antenna. For the initial form of the surface discharges of the average apparent charge equal to
qavg = 895 pC, the growth of the amplitude of the registered UHF pulses compared to the disk antenna by 7.80 ± 0.9 times on average was found. In turn, in case of high energy creeping sparks with an average apparent charge equal to
qavg = 5990 pC, lower growth of the amplitude, equal to 1.75 ± 0.7 times on average was observed. The comparison of the exemplary time waveforms of the PD pulses registered by the prototype antenna and commercial disk antenna is demonstrated in
Figure 11.
2.2.3. High-Frequency Current Transformer
The high-frequency current transformer is an inductive PD detector, which is usually installed on the wire grounding the investigated high voltage device. That wire is simultaneously a primary winding of the current transformer. A typical sensor HFCT consists of toroidal ferrite core, on which from several to over a dozen turns (secondary winding) are wound-up. This winding is terminated with a low-inductance sense resistor. A current pulse of high frequency flowing through the grounding wire (primary winding) causes the occurrence of a time-varying magnetic field, which passes through the secondary winding and induces a voltage across the sense resistor.
Figure 12 presents a model and equivalent circuit of the HFCT sensor, where
i is the current flowing through the primary winding,
R is the sense resistor,
VR an induced voltage in the secondary winding,
Lm is the magnetizing inductance of the current transformer,
Rc is the core parallel equivalent resistance,
Ll and
Rs are the leakage inductance and series resistance of the secondary winding, respectively. The remaining parasitic components included in the model are stray capacitance
C and inductance
LR of the sense resistor [
91].
The HFCT sensors are applied in the diagnostics of high-voltage power cables, power generators, metal-enclosed switchgears, and gas-insulated systems (GIS) [
92,
93,
94]. In the case of power transformers, the HFCT sensor may be installed on a grounding wire of the tank, on a grounded measuring tap of the bushing insulator, or a grounding of the neutral point of the transformer winding. The last option assures the highest detection sensitivity of the PDs occurring in the power transformer main insulation system.
The toroidal HFCT sensor core is most commonly made from soft ferrite. This material is ferromagnetic and consists of iron oxides mixed with oxides of manganese and zinc (MnZnFe
2O
4), or nickel and zinc (NiZnFe
2O
4). Manganese-zinc soft ferrites have high relative permeability (
µr ≈ 350–20,000 at 10 kHz), medium saturation flux density (
Bs ≈ 320−545 mT at 1200 A/m), operating frequency range from a few kilohertz to 4 MHz, and low resistivity
ρ (from 10 Ωm at 10 kHz to 0.01 Ωm at 100 MHz). In turn, nickel-zinc soft ferrites are characterized by low relative permeability (
µr ≈ 15−2000 at 10 kHz), slightly lower saturation flux density (
Bs ≈ 220−380 mT at 1200 A/m), much higher maximum operating frequency (up to about 200 MHz), and much higher resistivity
ρ (from 10
5 Ωm at 10 kHz to 10
3 Ωm at 100 MHz) [
95].
The clamp-type HFCT sensor devoted to PD monitoring in power transformers should have high sensitivity, wide bandwidth, closer to the constant line frequency response characteristics, and be equipped in enclosure shielding from external electromagnetic interferences. The sensitivity
S of the HFCT sensor depends on resistance value
R and the number of windings
N, as shown in Equation (2).
The lower −3 dB cutoff frequency of HFCT sensor is given by
where
Lm is the magnetizing inductance. The HFCT sensors are mostly loaded with non-inductive resistor
R = 50 Ω, in order to assure a good impedance matching with the transmission line (usually it is a standard 50-ohm coaxial cable) and with the analog input of signal acquisition unit. Therefore, to achieve a low value of the cutoff frequency
fL, the magnetizing inductance should be possibly the highest. The magnetizing inductance
Lm on the secondary side is given by
where
μ0 is the vacuum permeability (magnetic constant),
Ac is the core cross-sectional area, and
lc is the toroidal mean length. Equation (4) shows that the high value of the magnetizing inductance
Lm may be obtained the easiest using a large number of turns
N. Unfortunately, with increasing the number of turns, the sensor sensitivity is reduced. The other way is a choice of a core made of a material with high relative permeability
µr (e.g., MnZn soft ferrites) and possibly the smallest toroidal mean length
lc [
95,
96].
In turn, the higher –3 dB cutoff frequency of the HFCT sensor mainly depends on stray capacitance
C and the leakage inductance
Ll (Equation (5)). To obtain a high value of
fH, both parameters should be reduced.
The stray capacitance
C consists of several capacitive couplings, which are present in high-frequency current transformer, i.e., turn-to-turn capacitance, winding-to-core capacitance, winding-to-shielded enclosure capacitance, and primary-to-secondary winding capacitance. The stray capacitance can be minimized by reducing the number of turns
N, increasing the insulation layer of core and secondary winding (using a material with low dielectric constant
εr), increasing the distance between turns, and using a Faraday electrostatic shield [
95].
The reduction of the number of turns N is also favorable from the viewpoint of leakage inductance Ll minimizing. Additionally, the secondary winding should be made wide and have a small thickness.
As one may notice, the factors which have the greatest influence on the sensitivity and width of the HFCT sensor band, are the number of windings
N and the type of material, which the toroidal core is made from. The cores made from MnZn and NiZn soft ferrites were selected for the tests. The physical parameters of the materials are listed in
Table 4.
The dimensions of the applied ferrite cores were selected so that, after equipping them in shielding enclosure, they could be easily installed on the wire grounding the transformer tank or the neutral point of the star windings. Since the width of the grounding wire/rail of power transformer usually does not exceed 40 mm, it was decided to choose toroidal cores with an internal diameter equal to 65 mm.
The external diameter MnZn ferrite core was 107 mm, and its height was 18 mm. For NiZn ferrite core they were, respectively, 103 and 16 mm. To reduce the skin effect, for manufacturing of the secondary winding, the high-frequency Litz-wire with a diameter of 0.48 mm and external insulation with a thickness of 0.02 mm made from natural silk of dielectric constant εr = 2.6 was used.
Litz-wire consists of 920 strands insulated electrically from each other, wherein a diameter of a single wire is 0.016 mm (AWG54). The cores were split in half using a diamond cutting disc. The secondary winding with a counter-wound compensation turn was applied to one of the core halves. To reduce the stray capacitance (or more precisely, the turn-to-turn capacitances), the distance between individual turns was maximized. The enclosure of HFCT sensor was made from aluminum with a thickness of 2 mm to attenuate radiofrequency electromagnetic interferences and improved performance in an electrically noisy substation environment. It was equipped with a 1 mm slit for better magnetic coupling between primary and secondary winding [
97]. The sensor HFCT housing and the housing of the conditioning and A/D processing module were integrated and jointly are a complete sensor ISO-002-HF with the function of fiber optic signal transmission (
Figure 13).
The investigation on the influence of the number of turns
N and material of the used core on the frequency response of the prototype HFCT sensors was performed using the measurement setup presented in
Figure 14. The measurement setup consisted of the oscilloscope Tektronix MDO 3104 with the sampling frequency of 5 GS/s, 1 GHz bandwidth, and built-in signal generator. The sinusoidal signal was generated in the frequency range from 100 kHz to 30 MHz with a step frequency of 100 kHz. It allowed to cover the full range of high-frequency (HF) band, i.e., 3–30 MHz. The current flow in the primary winding was forced by using a 50 Ω noninductive resistor
RL in series. The same resistor was connected in parallel with the secondary winding of the HFCT sensor.
The measurement of the frequency response was automatically realized by the application implemented in LabVIEW environment and launched on a computer connected to the oscilloscope. The individual points of the frequency response curve
FRHFCT(f) were calculated as
where
VR(
f) is the RMS value of voltage measured on the output of the HFCT sensor at frequency
f,
VGEN(
f) is the RMS value of voltage measured on the output of signal generator at frequency
f.
Figure 15 presents the frequency response curves of the prototype HFCT sensors. As one may notice, simultaneously with the increase in the number of turns, the sensor sensitivity decreases. It results from the fact that the values of that parameter depend directly proportional to the load resistance
R (which in this case was constant and equal to 50 Ω) and inversely proportional to the number of turns
N. Increasing the number of turns raises the magnetizing inductance
Lm, which causes the reduction of lower −3 dB cutoff frequency
fL of the HFCT sensor. This effect is even more visible for sensors with MnZn soft ferrite core, whose magnetizing inductance is high regarding the high value of permeability
µ at low frequencies. For comparison, the value
fL for current transformer with MnZn ferrite core and 10 turns is lower than 100 kHz, in turn, for a current transformer with NiZn ferrite core with the same number of turns,
fL equals 450 kHz. For a lower number of turns, the differences are even more clear. For instance,
fL for HFCT sensor with MnZn ferrite core and
N = 3 is also very low and equals 250 kHz. The current transformer of the same construction, but with the ferrite core NiZn, has
fL equal to even 3 MHz. It should be emphasized that for the HFCT sensors devoted to PD detection, the more important parameter than the
fL is higher −3 dB cutoff frequency
fH, since the discharges in oil-paper insulation emit high-frequency waves mainly in the range between approx. 2 and 20 MHz [
98]. In this case, it is more favorable to apply a lower number of turns. It allows to obtain high sensitivity
S of the HFCT sensor and high value of the parameter
fH, which improves the performance of the sensor in the range of higher frequencies (> 1 MHz). One may notice in
Figure 14, that the positive effect resulting from reduction of the number of turns is more for sensors with the core made from NiZn soft ferrite since permeability
µ of the applied material is a practically constant value in the wide range between 1 MHz and approx. 50 MHz.
Based on the analysis of the measured frequency response curves, for further research, the two prototype HFCT sensors were selected. The first of the chosen sensors was manufactured from ferrite core MnZn with the three-turn secondary winding. The main advantages of that construction are a very high sensitivity (higher than popular commercial sensors), wide bandwidth (250 kHz–25.5 MHz), and relatively flat frequency response curve. The second from the selected constructions was built from ferrite core NiZn with four turns. It is characterized by slightly lower sensitivity (≤ 2.67 dB) than the transformer with MnZn core, but its frequency response curve is almost ideally flat in a wide band between 1.8 and 30 MHz. In the first stage of the laboratory tests, measurements of the impulse response of the selected prototype HFCT sensors were conducted. For this purpose, a standard LDC-5 PD calibrator (dDoble Engineering Company, Marlborough, USA) was used, which allowed to inject pulses with an apparent charge of 500 pC and duration of 50 ns into the primary circuit. The waveforms of the calibration pulses induced in the secondary circuit of the HFCT sensor were registered using an oscilloscope (
Figure 16).
Figure 17 shows the recorded waveforms of the impulse response of the tested HFCT sensors. The highest detection sensitivity of the pulse of apparent charge 500 pC was shown by the prototype current transformer made from MnZn ferrite core with three turns. At the output of this current transformer, the pulses of amplitude
Amax = 146 mV were registered, which means that its sensitivity was by about 70% higher than the prototype sensor made from NiZn ferrite core (
Amax = 86 mV) and commercial HFCT sensor type RFCT-4 (
Amax = 88 mV).
Time parameters of the pulses generated by standard PD calibrator are optimized for the conventional method IEC 60270, in which case the registration of PD pulses is performed in a relatively low frequency band (100–500 kHz). For that reason, the tests with the use of the PD calibrator do not allow unambiguous statement that the prototype HFCT sensor with core MnZn is an optimal construction of the detector of high-frequency PD pulses. Nevertheless, measurements with the use of calibrator showed that due to the lowest lower −3 dB cutoff frequency (fL = 250 kHz) among the tested HFCT sensors, it may be successfully employed both in the standardized frequency range according to IEC 60270 as well as at higher frequencies up to 25.5 MHz.
In the second stage of the laboratory research, the detection sensitivity of PD pulses generated in oil-paper insulation was tested, which were simultaneously registered by four HFCT sensors (two prototype and two commercial). The research was conducted with the use of a transformer tank filled with mineral oil. The electrode systems devoted to generating PD in oil and surface PD on the pressboard sample were put inside the tank. Detailed information concerning the geometry of the applied electrode systems was presented in [
24]. The HFCT sensors were installed on grounding wire going out of the tank through the ceramic bushing insulator. The PD pulses were registered with the use of a four-channel oscilloscope Tektronix MDO3104 with the sampling frequency of 5 GS/s. Additionally, the standard IEC 60270 measurement setup was applied to control the level of the PD apparent charge (
Figure 18). Discharges in oil of apparent charge in the range between about 200 and 850 pC were generated at 29 kV. In turn, the surface discharges had a significantly higher intensity and apparent charge (
q = 450–5100 pC). This type of discharge was generated by a voltage of 15.4 kV.
Figure 19 presents typical partial discharge time waveforms and power spectral density (PSD) characteristics registered with the use of HFCT sensors. The results of frequency analysis showed that PDs in oil generated pulses of high frequency in the band between 12.4 and 18.3 MHz, whereas surface PD generated pulses in a narrow band between 10.8 and 11.6 MHz. The PD detection sensitivity of both prototypes was higher than commercial sensors. In the case of discharges in oil, the highest sensitivity was observed for the prototype HFCT sensor with four turns and NiZn soft ferrite core. Its sensitivity was 36% higher on average than the sensors from MnZn soft ferrite with three turns. On the other hand, the construction based on the MnZn core showed higher sensitivity in the detection of surface discharges on the pressboard sample. The average growth in sensitivity compared to the prototype with NiZn core was 46%. The detection possibility of surface discharges is very important from the viewpoint of the performance and effectiveness of the monitoring system operation. It is one of the most energetic and destructive types of partial discharges. Therefore, for the designed monitoring system, finally, the HFCT sensor with MnZn core was selected.